This paper presents the analysis and design of an active electromagnetic interference (EMI) filter (AEF) for the commonmode (CM) noise reduction of switching power converters. The features of the several types of AEFs are discussed and compared in terms of implementation. The feedforward AEF with a voltagesensing and voltagecancellation (VSVC) structure is implemented for an LLC resonant converter to replace a multiplestage passive EMI filter and thereby reduce CM noise. The characteristics and performance of the VSVCtype AEF are investigated through theoretical and experimental works.
I. INTRODUCTION
The reduction of electromagnetic interference (EMI) is a significant issue in the design and implementation of highpower density switching power converters. A powerline EMI filter is a traditional solution to reduce the conducted emissions of converters
[1]
. A passive filter with a commonmode (CM) choke and X and Y capacitors is generally used for this purpose. However, this filter is bulky, and the manufacturing of the filter components, such as the CM choke, is too laborious.
In recent years, active EMI filters (AEFs) have been considered as an alternative to passive filters. AEFs perform an active cancellation of noise instead of LC filtering. Hence, they do not require bulky passive components. Moreover, they show great potential for integration into EMI filters in a small chip or package.
Research on AEF can be found in the literature
[2]

[10]
. The active cancellation of noise currents using AEFs was studied in
[2]

[7]
. Voltage cancellation and hybrid methods were reported in
[8]

[10]
. Differential mode (DM) AEFs were also considered in
[5]
,
[10]
. Despite the extensive study on the usefulness of AEFs, further research on AEF circuits and components should still be performed for practical design and implementation.
This paper presents the design and implementation of active EMI filters for the CM noise reduction of LLC resonant converters. The characteristics of several AEF topologies are discussed and compared. The control loop characteristics of feedback and feedforward AEFs are also investigated. A feedforward AEF with a voltagesensing and voltagecancellation (VSVC) structure is finally implemented to replace the multiplestage passive EMI filter. The performance of the implemented AEF in terms of EMI reduction is investigated through experimental works.
II. STRUCTURE OF ACTIVE EMI FILTER
 A. Active EMI Filter Topologies
Fig. 1
shows the structure of the AEF for the CM noise reduction of switching power converters. The conducted noise of the power line is measured and is actively cancelled by the compensating voltage or current generated from the control amplifier.
Structure of active EMI filter for CM noise reduction.
Four types of AEF topologies can be considered for the combination of the sensing and cancellation methods. The simplified CM equivalent circuits of four AEF types are shown in
Fig. 2
. In terms of implementation, voltage sensing is simpler than current sensing because it does not require a current transformer. Meanwhile, the current capability of control amplifiers is crucial for the selection of a cancellation method. The current type is difficult to implement because of the lack of a highcurrent and widebandwidth operational amplifier (OP amp). The possibility of a high CM current occurring is another problem for this method because it makes a bypassing path to the ground. As a result, the VSVC type is considered for the implementation of the AEF in this study.
Types of AEFs.
 B. Characteristics of Feedback and Feedforward AEFs
In view of the control loop, two AEF structures can be considered according to the point of the measurement of noise signals. A noise signal is measured at the LISN terminals (control target) in the feedback control and at the converter input (noise source) in the feedforward control.
Fig. 3
shows the CM equivalent circuits of the VSVCtype feedback and feedforward AEFs. The transfer functions of both AEFs are derived to investigate the characteristics of the control loops. The transfer function of the feedback AEF is given as
where
V_{s}
,
V_{o}
,
Z_{s}
,
R_{L}
, and
A
denote the voltage of the noise source, voltage of the LISN terminal, noise source impedance, resistance of the LISN, and forward gain of the control loop, respectively. The measurement of noise source impedance is discussed in
[11]
. The total impedance
Z_{L}
of the LISN side can be represented as
or
for the circuits with or without the additional CM choke, respectively, where
L_{CM}
is the inductance of the additional CM choke. The small CM choke is generally combined to reduce the EMI in the high frequency range over few tens MHz because of the frequency limitation of the filter components, such as the OP amp and injection transformer.
CM equivalent circuit of feedback and feedforward AEFs. (a) Feedback AEF (b) Feedforward AEF.
The transfer function of the feedforward AEF can also be derived from
Fig. 3
(b) as
As indicated in (1), an extremely high value of the forward gain
A
is required to minimize the noise voltage (
V_{o}
) at the LISN terminal for the feedback AEF. However, the noise voltage tends to drop to zero as the forward gain
A
approaches unity (
A
= 1) for the feedforward AEF, as shown in (4). Therefore, the feedforward structure is chosen for the implementation.
III. DESIGN AND IMPLEMENTATION
The CM circuit diagram of the VSVC type feedforward AEF is shown in
Fig. 4
. The diagram consists of a sensing circuit, a control amplifier, and a voltage injection transformer. The forward gain
A
can be represented as follows using the three blocks:
where
G_{T}
,
G_{c}
, and
H_{s}
are the transfer functions of the injection transformer, control amplifier, and sensing circuit, respectively. The characteristics and design of each part are discussed in the next section.
CM circuit of VSVCtype feedforward AEF.
 A. Sensing Circuit
The sensing circuit in
Fig. 4
is a firstorder highpass filter. The transfer function of this circuit can be given as follows using the concept of virtual ground:
Its cutoff frequency is given as
f_{h}
=1 / (2
π
R
_{1}
C_{s}
) . The cutoff frequency of the sensing circuit should be lower than the lowest frequency component of the noise voltage. It is thus determined to be the value that is sufficiently lower than the lowest switching frequency
f_{s}
of the LLC resonant converter (
f_{s}
= 80
kHz
in this design).
Fig. 5
shows the simulated result for the frequency response of the sensing circuit, where
R
_{1}
=10
k
Ω ,
C_{s}
= 20
nF
, and
f_{h}
= 796
Hz
.
Simulated frequency response of sensing circuit ( R_{1} =10 kΩ , C_{s} = 20 nF and f_{h} = 796 Hz ).
 B. Control Amplifier
The OP amp is generally used for a control amplifier that generates a cancellation voltage. The gain of the inverting amplifier shown in
Fig. 4
can be represented as
where
R
_{1}
,
R
_{2}
, and
ω_{c}
( = 2
π
f_{c}
) are the feedback resistor, input resistor, and −3 dB frequency of the OP amp, respectively. The phase response of the OP amp circuit is more important than the gain because the phase delay of the control amplifier severely degrades the performance of the feedforward AEF, especially in the high frequency range. In the worst case, a large phase delay may amplify the noise by summing up the noise and cancellation voltages.
Fig. 6
shows the measured gain/phase responses of the inverting amplifier using the OP amp OPA847, where
f_{c}
= 22
MHz
. Several important parameters of OPA847 are summarized in
Table 1
. The frequency response of the OP amp in the low gain (
G_{c}
 < 1) is important for the high frequency range, but it is not a major concern in practice because the injection transformer exhibits a relatively slow response, which is dominant in the control loop.
Measured gain/phase response G_{c}(s) of inverting amplifier using OPA847 ( R_{1} =10kΩ , R_{2} = 5kΩ ).
OP AMP SPECIFICATIONS (OPA847)
OP AMP SPECIFICATIONS (OPA847)
The output voltage swing and current capability of the OP amp are critical parameters in real implementation. The OP amp should supply sufficient current to magnetize the injection transformer, and the input voltage of the transformer should be within the maximum output voltage swing
V_{op,max}
. These requirements are discussed in the next section.
 C. Injection Transformer
The output voltage of the OP amp is applied to the primary terminal of the transformer, and the CM noise voltage
V_{n}
can be canceled by the secondary voltage
V_{c}
connected in series to the CM path. The CM equivalent circuit of the transformer is shown in
Fig. 7
(a); its simplified form is shown in
Fig. 7
(b)
[12]
, where
and
n
=
N_{s}
/
N_{p}
. The symbols
C
_{1}
,
C
_{2}
,
C
_{12}
,
L
_{l1}
,
L
_{l2}
, and
L_{m}
denote the capacitance of the primary terminal, capacitance of the secondary terminal, interwinding capacitance, leakage inductance of the primary winding, leakage inductance of the secondary winding, and magnetizing inductance, respectively.
Circuit model of injection transformer. (a) CM equivalent circuit model. (b) Simplified model.
The transfer function of the injection transformer can be derived from
Fig. 7
(b) and for
L_{m}
>>
L_{lk}
as
where
As shown in (11) and (12), the leakage inductance
L_{lk}
and equivalent winding capacitance
C_{eq}
are the parameters that determine the bandwidth of the injection transformer. Thus, these parameters should be minimized to reduce high frequency noise.
The maximum output voltage swing and current capability of the OP amp should also be considered in the transformer design. The peak value of the primary voltage
V_{p, peak}
should be within the maximum output swing
V
_{op,max}
of the OP amp; that is,
where
V_{s,peak}
denotes the peak voltage of the transformer output. The input current of the transformer is also limited by the maximum output current
I
_{op,max}
of the OP amp, as shown in (14).
where
Z_{Ti}
denotes the input impedance of the transformer. The primary turns of the transformer can be determined as
where
and
μ
_{0}
,
μ_{r}
,
A_{c}
, and
l_{c}
denote air permeability, relative permeability, crosssectional area, and mean length of the magnetic core, respectively.
The selection of core materials is extremely important in miniaturizing transformer size and improving high frequency characteristics. The large number of turns of the transformer winding increases the leakage inductance and winding capacitance, which in turn degrade the frequency response. Hence, the core material should exhibit high permeability to reduce winding turns. A nanocrystalline core with
μ_{r}
= 14,000 is used for the implementation. The parameters of the implemented injection transformer are listed in
Table II
.
PARAMETERS OF IMPLEMENTED TRANSFORMER
PARAMETERS OF IMPLEMENTED TRANSFORMER
Fig. 8
shows the measured gain/phase response
G_{T}
(
s
) of the injection transformer, the poles of which are located at 10.9 MHz. The measured total response
G_{c}
(
s
)
G_{T}
(s), including the OP amp and transformer, is shown in
Fig. 9
. Both figures show that the frequency response of the control loop is governed by the transformer characteristics.
Measured gain/phase response of the transformer G_{T}(s).
Measured gain/phase response of the OP amp and transformer G_{c}(s)G_{T}(s).
 D. Discussion on Stability
The feedforward AEF does not have any feedback signals in the control loop. However, we note that in (4), the natural feedback loop is made by the impedance of the noise source
Z_{s}
. We can rewrite (4) for the voltage
V_{L}
shown in
Fig. 4
as
If the source impedance
Z_{s}
= 0, then the transfer function is (1 –
A
), and only the feedforward loop exists. However, for the case in which
Z_{s}
≠ 0 , a feedback loop is made with the forward and feedback gains of (1 –
A
) and
Z_{s}
/
Z_{L}
, respectively. Consequently, the operation of the AEF becomes unstable for a certain condition of gain
A
.
Figs. 10
and
11
show the Bode diagrams of the loop gain (1 −
A
)·
Z_{s}
/
Z_{L}
for the DC gain of
G_{c}
(
s
)
G_{T}
(
s
), (
R
_{2}
/
R
_{1}
)·
n
= 0.98 and 1, respectively, where
Z_{s}
=1/
j
(2
πf
·1
nF
) ,
C_{L}
= 0.1 uF, and
R_{L}
= 50 Ω. As shown in
Fig. 10
, the gain and phase margins are sufficient, and the feedback loop is stable. However, these margins decline rapidly for (
R
_{2}
/
R
_{1}
)·
n
= 1, as shown in
Fig. 11
. Therefore, this condition should be avoided for the stable operation of the feedforward AEF.
Bode plot of loop gain (1 − A)Z_{s}/Z_{L} for n(R_{2}/R_{1}) = 0.98.
Bode plot of loop gain (1 − A)Z_{s}/Z_{L} for n(R_{2}/R_{1}) = 1.
IV. EXPERIMENTAL RESULTS AND DISCUSSION
Figs. 12
and
13
show the implemented AEF and experimental setup, respectively. The LLC resonant converter with a power rating of 500 W is used for the test, in which the AEF replaces the threestage passive CM filter. The tested conditions are given in
Table III
.
Experimental setup.
Photograph of implemented AEF.
EXPERIMENTAL CONDITIONS
Fig. 14
shows the CM noise spectrum without any EMI filter, with the reference line being the EN55022 limit. The CM noise is severe in the frequency range of 150 kHz to 3 MHz. The CM noise spectrum with the threestage passive filter is shown in
Fig. 15
, in which the CM noise is below the limit line.
CM noise spectrum without an EMI filter.
CM noise spectrum with a threestage passive EMI filter.
Fig. 16
shows the experimental waveforms of the AEF and CM noise spectrum for (
R
_{2}
/
R
_{1}
)·
n
= 1, where
R
_{1}
=10
k
Ω and
R
_{2}
= 5
k
Ω . As shown in
Fig. 16
(a), the output voltage
V_{op}
of the OP amp is saturated to
V_{op,max}
and cannot fully cancel the noise voltage. The low frequency spectrum exceeds the desired EMI limit, as shown in
Fig. 16
(b). As discussed in the previous section, the feedback loop is unstable for this condition, and the output voltage of the OP amp is saturated. This problem can be solved by applying the slightly reduced OP amp gain.
Experimental results for (R_{2}/R_{1})·n = 1, R_{1} =10kΩ , and R_{2} = 5kΩ . (a) Experimental waveforms of AEF. (b) CM noise spectrum.
Fig. 17
shows the experimental results for (
R
_{2}
/
R
_{1}
)·
n
= 0.97, where
R
_{1}
=10.3
k
Ω and
R
_{2}
= 5
k
Ω . The figure clearly shows the significant improvement in the EMI performance. The peak output voltage of the OP amp is within the maximum voltage swing
V_{op,max}
, and the sensed noise voltage
V_{ns}
and OP amp output
V_{op}
exhibit nearly the same shape, as shown in
Fig. 17
(a). Thus, the CM noise voltage is successfully canceled, and an extremely small LISN voltage
V_{o}
is observed. The CM noise spectrum for this condition is shown in
Fig. 17
(b). The noise margin of over 10 dB for the EN55022 limit line can be obtained for the frequency range of 150 kHz–20 MHz. As predicted in the frequency response of
G_{c}
(
s
)
G_{T}
(
s
) shown in
Fig. 10
, the AEF performance declines at a frequency of over 15 MHz because of the limited bandwidth of the injection transformer. In the frequency range below 20 MHz, the performance of the implemented AEF is comparable to that of the threestage passive filter shown in
Fig. 15
. A noise voltage above this range can be reduced by employing a small highfrequency CM choke.
Experimental results for (R_{2}/R_{1})·n = 1, R_{1} =10.3kΩ , and R_{2} = 5kΩ . (a) Experimental waveforms of AEF. (b) CM noise spectrum.
V. CONCLUSIONS
This study presented the design and implementation of an AEF for the CM noise reduction of switching power converters. A VSVCtype feedforward AEF was considered for the implementation and used to replace the threestage passive EMI filter. The practical considerations for the implementation of the filter were provided. Such considerations include the output voltage swing and current capability of the OP amp, the frequency characteristics of the filter components, and the stability of the feedforward AEF. The operation and performance of the implemented AEF were investigated through the experimental works for an LLC resonant converter with a power rating of 500 W. The experimental results show that a noise margin of over 10 dB for the EN55022 limit can be obtained for the frequency range of 150 kHz–20 MHz by employing the implemented AEF. The designed AEF can replace bulky passive filters and be successfully used to reduce lowfrequency EMI.
BIO
KukHee Lee was born in Busan, Korea, in 1992. She received her B.S. and M.S. degrees in Control and Instrumentation Engineering from Gyeongsang National University, Jinju, Korea, in 2014 and 2016, respectively. Her research interests include switching power converters and magnetic design.
ByeongGeuk Kang was born in Hapcheon, Korea, in 1983. He received his B.S. and M.S. degrees in Control and Instrumentation Engineering from Gyeongsang National University, Jinju, Korea, in 2008 and 2010 respectively. He is now working toward his Ph.D. degree at the same university. His research interests include switching power converters and magnetic design.
Yongoh Choi was born in Miryang, Korea, in 1986. He received his B.S. and M.S. degrees in Control and Instrumentation Engineering from Gyeonsang National University, Jinju, Korea, in 2011 and 2014, respectively. His research interests include switching power converters and wireless power transform.
SeKyo Chung was born in Daegu, Korea, in 1966. He received his B.S. degree in Electronic Engineering from Kyungpook National University, Daegu, in 1989, and his M.S. and Ph.D. degrees in Electrical Engineering from the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, in 1992 and 1997, respectively. Since 1997, he has been with the Department of Control and Instrumentation Engineering, Gyeongsang National University, Jinju, Korea, where he is a Professor and Principal Researcher of the Engineering Research Institute. He was a Visiting Professor of Electric Power Control Lab, Kyushu Institute of Technology, Kitakyushu, Japan, in 2000, and a Visiting Scholar at Texas A&M University, College Station, TX, USA, in 2002 and 2012. His research interests lie in the areas of power electronics and control, which include highperformance ac machine drives, switching power converters, magnetic design, and power electronics interface to microgrids. Dr. Chung is an Associate Editor of Journal of Power Electronics.
JaeSun Won was born in Wonju, Korea, in 1973. He received his M.E. and Ph.D. degrees in Electrical Engineering from Yeungnam University, Gyeongsan, Korea, in 1999 and 2005, respectively. He is currently working with the Wireless Power Transfer (WPT) Team of the Digital Module Division of Samsung ElectroMechanics, Suwon, Korea, as a Principal Engineer. His research interests include resonant inverter/converter systems, soft switching technology, electromagnetic interference (EMI) design of power electronics, and wireless power transfer systems.
HeeSeung Kim was born in Seoul, Korea, in 1982. He received his B.S., M.S., and Ph.D. degrees in Electronic Engineering from Kookmin University, Seoul, Korea, in 2008, 2010, and 2013, respectively. He is currently working with the Wireless Power Transfer (WPT) Team of the Digital Module Division of Samsung ElectroMechanics, Suwon, Korea, as a Senior Engineer. His research interests include magnetic devices, electromagnetics analysis, power conversion system design, electromagnetic interference (EMI) design of power electronics, and wireless power transfer systems.
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