DC microgrids are considered as prospective systems because of their easy connection of distributed energy resources (DERs) and electric vehicles (EVs), reduction of conversion loss between dc output sources and loads, lack of reactive power issues, etc. These features make them very suitable for future industrial and commercial buildings’ power systems. In addition, the bipolartype dc system structure is more popular, because it provides two voltage levels for different power converters and loads. To keep voltage balanced in such a dc system, a bidirectional dual buckboost voltage balancer with direct coupling is introduced based on Pcell and Ncell concepts. This results in greatly enhanced system reliability thanks to no shootthrough problems and lower switching losses with the help of power MOSFETs. In order to increase system efficiency and reliability, a novel burstmode control strategy is proposed for the dual buckboost voltage balancer. The basic operating principle, the current relations, and a smallsignal model of the voltage balancer are analyzed under the burstmode control scheme in detail. Finally, simulation experiments are performed and a laboratory unit with a 5kW unbalanced ability is constructed to verify the viability of the bidirectional dual buckboost voltage balancer under the proposed burstmode control scheme in lowvoltage bipolartype dc microgrids.
I. INTRODUCTION
DC microgrids
[1]

[14]
, compared with conventional ac microgrids, have attracted a lot of attention in the pursuit of effective solutions to meet modern energy distribution challenges. They can provide a better connection for dc output type sources such as photovoltaic (PV) systems, fuel cells, and energy storage devices (Liion batteries and supercapacitors, etc.). Moreover, system efficiency becomes higher for the lower conversion losses of the inverters between sources and loads, when loads are directly supplied with dc power. Other advantages are as follows: (1) no frequency stability or reactive power issues; (2) no skin effect or ac losses; (3) no downtime related to voltage sags or blackouts. Thus, they are very suitable for future industrial and commercial buildings’ power systems, and for making our daily life efficient and secure.
Generally, there are two types of dc microgrids
[1]
,
[4]
: unipolartype and bipolartype. The first type has only one voltage level in a twowire dc distribution system. This makes it impossible to supply some types of loads at half voltage. The other type has two voltage levels in a threewire system. This can provide a grounded neutral line, which is favorable for the security of people in domestic and office places. Additionally, bipolartype systems have higher reliability due to the possibility of using a power supply under one line failure. This can also decrease power loss for reducing current levels and increasing power transmission capability.
As shown in
Fig. 1
, a lowvoltage bipolartype dc microgrid
[1]
,
[11]
consists of a bidirectional rectifier, a voltage balancer, distributed energy sources (DERs) and dc/ac loads, where dc power is transmitted through a threewire system. The bidirectional rectifier can construct a unipolartype system, and then a voltage balancer with the voltage balancing function is introduced to build a threewire bipolartype system. The voltage balancer can make the power supply flexible and reliable with two voltage levels for different loads in a bipolartype dc microgrid. Compared with the unipolartype dc grid, a voltage balancer is needed and additional loss can be produced in a bipolartype dc grid.
Lowvoltage bipolartype dc microgrid structure.
This paper will focus on the research of a voltage balancer. The conventional topologies of bridgetype converters can suffer from a shootthrough risk, which is a major drawback to the reliability of this type of voltage balancer
[4]
. In addition, MOSFETs with lower switching loss and conducting loss cannot be used directly in conventional bridge converters due to the poorer characteristic of the MOSFET bodydiode. Thus, a bidirectional dual buckboost voltage balancer based on the Pcell and Ncell concepts is adopted in this paper
[15]

[18]
. Compared with the traditional bridgetype converter, it exhibits the following distinct merits
[19]

[24]
: (1) there is no shootthrough issue due to the fact that there are no active power switches connected in series in each phase leg; (2) the reverse recovery dissipation of the power switch is greatly reduced because there is no freewheeling current flowing through the body diode of the power switches which improves the reliability; (3) using MOSFETs combined with optimized diodes permits a higher switching frequency to reduce weight and volume. However, the bidirectional dual buckboost voltage balancer has the disadvantage that two separate inductors are needed resulting in increased volume and weight. However, this can be mitigated by using the directly coupled inductor technique.
Generally, a voltage balancer is always working under constantvoltage mode control, resulting in additional power loss, such as converters’ switching loss and conduction loss, especially under voltage balanced conditions. To increase system efficiency and reliability, a novel burstmode control strategy is proposed for the dual buckboost voltage balancer.
In this paper, a bidirectional dual buckboost voltage balancer based on the Pcell and Ncell concepts is introduced in Section II, where a direct coupling inductor combined with a common inductor is presented to optimize the two separate inductors. The basic operation principle based on the continuous conduction mode (CCM) is also presented and analyzed in this section. In Section III, a novel burstmode control strategy is proposed, where the operating principle and the current relations of the inductors, the capacitors and the unbalanced loads are analyzed in detail. In Section IV, an average smallsignal model of the voltage balancer under the CCM condition is given to design the control system parameters. In Section V, a 10kW simulation experiment is carried out to validate the feasibility of the proposed burstmode control strategy for the voltage balancer. In Section VI, a laboratory unit with a 5kW unbalanced ability is constructed to verify the viability of the proposed burstmode control strategy for the dual buckboost voltage balancer in lowvoltage bipolartype dc microgrids. Finally, some concluding remarks are drawn in Section VII.
II. BIDIRECTIONAL DUAL BUCKBOOST VOLTAGE BALANCER WITH DIRECT COUPLING
The concept of a “lowvoltage bipolartype dc microgrid” with a higher power supply quality is shown in
Fig. 1
, where medium voltage is converted into dc voltage in a range from 360V to 400V through a stepdown transformer and an activefrontend rectifier. The threewire dc distribution consisting of a positive line, a neutral line and a negative line is constructed by the voltage balancer, which provides two different voltage levels for customers. The loadside dcdc/dcac converters can be connected between the positive line and the negative line, the positive line and the neutral line, the negative line and the neutral line separately. Since the latter two connections may cause power and voltage imbalances between the positiveneutral and the negativeneutral lines, it is essential to adopt dc voltage balance control. The voltage balancer is placed near the bidirectional rectifier to balance the positive and negative voltage, which can also be placed near the load side.
 A. Bidirectional Dual BuckBoost Voltage Balancer Based on Pcell and Ncell
Power electronics circuits can be constructed with two basic switching cells defined as Pcell and Ncell
[16]
[17]
, as shown in
Fig. 2
. Each cell consists of one switching device (a MOSFET, IGBT or any other switching device) and one diode constituting three terminals: (+) which is connected to the positive of a voltagesource or capacitor, () which is connected to the negative of a voltagesource or capacitor, and a common terminal which is shown as (→) or (←). For the Pcell, this common terminal is connected to the negative terminal of a current source or inductor. For the Ncell, the common terminal is connected to the positive of a current source or inductor.
Two basic switching cells: Pcell and Ncell.
Fig. 3
shows the buckboost converters based on the basic switch cells. It can be seen that the Pcell buckboost converter can only transfer power from upside to downside, and the Ncell buckboost converter from downside to upside. Thus, by combing the former two converters, a bidirectional dual buckboost converter can be obtained forming a voltage balancer, as shown in
Fig. 4
. The voltage balancer consists of the Pcell leg (
S
_{1}
,
D
_{1}
), the Ncell leg (
S
_{2}
,
D
_{2}
), separate inductors (
L
_{1}
,
L
_{2}
), and two split dc bus capacitors (
C_{u}
,
C_{d}
). MOSFETs are used as power switches here, and their body diodes never work. Thus, both the power switches and the power diodes can get an optimized design independently. This permits a higher switching frequency to reduce the weight and volume.
Buckboost converters based on basic switching cells.
Bidirectional dual Buckboost voltage balancer.
As shown in
Fig. 4
,
V_{dc1}
,
V_{dc2}
and
V_{dc}
stand for the positiveneutral, negativeneutral and positivenegative voltages, respectively. In the following analysis, it is assumed that
V_{dc}
is kept constant with the regulation by the bidirectional rectifier.
 B. Mathematical Model in the Rotor Reference Frame
To overcome the drawback of the bidirectional dual buckboost converter, where two separate inductors are needed resulting in more volume and weight, the directly coupled inductor technique combined with a common inductor is adopted, as shown in
Fig. 5
. The equivalent inductance
L_{pn}
is four times each phase inductance (
L_{d1}
,
L_{d2}
) using the direct coupling technique. The inductor
L_{pn}
between
S
_{1}
and
S
_{2}
, and the stray inductance
L_{S}
of the interconnections will limit the current
i_{pn}
if there is any overlap in the switching of the Pcell and Ncell devices.
The directly coupled inductor technique combined with the common inductor.
It is assumed that the overlap time is limited to below some value Δ
i
_{limit}
. Then,
L
_{pn}
has the following relation:
Depending on the system requirements, the directly coupled inductor can be optimized using Eq. (1).
Based on the current ripple analysis, the total inductance value (
L_{d1}
+
L_{com}
) can be obtained as:
where,
f_{s}
is the switching frequency and
V_{dc1}
=
V_{dc2}
. From Eq. (1),
L_{d1}
(
L_{d2}
) can be derived, and
L_{com}
can also be calculated under the condition that Δ
I_{LD}
≤ 20%
I
_{L0}
(
I
_{L0}
nominal operating current).
 C. Basic Operation Principle under the CCM Condition
To simplify the analysis of the operation principle, some assumptions are made as follows: (1) all of the power MOSFETs and diodes are ideal devices with no consideration of the switching time or conduction voltage drop; (2) all of the inductors and capacitors are ideal with
L_{d1}
=
L_{d2}
and
C_{u}
=
C_{d}
=
C
; (3) the output dc voltages (
V_{dc1}
,
V_{dc2}
) are constant during each switching process.
The reference direction of
i_{L}
is from the p terminal to the common terminal o. It can be seen that
i_{L1}
is as a positive current while
i_{L2}
as a negative current.
Fig. 6
shows the four basic operation modes under the CCM condition.
Basic operation modes: (a) Positive current i_{L} = i_{L1}, S_{1} turned on; (b) Positive current i_{L} = i_{L1}, D_{1} freewheeling; (c) Negative current i_{L} = i_{L2}, S_{2} turned on; (d) Negative current i_{L} = i_{L2}, D_{2} freewheeling.
1) Mode 1:
When
R_{1}
is bigger than
R_{2}
, the voltage
V_{dc1}
of the capacitor
C_{u}
will increase combined with a decrease of the voltage
V_{dc2}
of the capacitor
C_{d}
. At this moment,
S
_{1}
is turned on and the positive inductor current
i_{L1}
increases linearly. In this mode, the Pcell leg starts to transfer additional power to
R_{2}
through
L
_{1}
, as shown in
Fig. 2
(a).
2) Mode 2:
When
S
_{1}
is turned off,
i
_{L1}
continues to run through the freewheeling diode
D
_{1}
, as shown in
Fig. 2
(b). The process is not be terminated until
S
_{1}
is turned on again.
3) Mode 3 and Mode 4:
Conversely, when
R_{1}
is smaller than
R_{2}
,
S
_{2}
is turned on and the negative current
i_{L2}
increases linearly. At this time, the Ncell leg starts to transfer the redundant power to
R_{1}
through
L
_{2}
, as shown in
Fig. 2
(c).
Fig. 2
(d) shows that
i_{L2}
continues to run through the freewheeling diode
D
_{1}
when
S
_{2}
is turned off.
III. THE NOVEL BURSTMODE CONTROL SCHEME FOR A BIDIRECTIONAL DUAL BUCKBOOST VOLTAGE BALANCER
Burst mode control has been widely used in PWM converters to achieve lower standby losses
[25]
,
[26]
. It is an operation mode using the cycleskipping technique to reduce the switching loss in a switching regulator and to increase the operating efficiency at low power levels. During a burst cycle, each burst of switching can be started at the lowest output voltage value (which occurs just at the start of a burst) and stopped at the maximum voltage value (which occurs at the end of a burst), and this process is repeated. This operation mode can result in high efficiency at low power levels, because the MOSFETs converters can operate at a designed power level close to the optimum efficiency point during a switching period.
Based on the idea of burst mode, a novel adaptive burst mode control scheme is proposed for the bidirectional dual buckboost voltage balancer in lowvoltage bipolartype dc microgrids.
Fig. 7
shows a simplified system control block diagram of this scheme. In the proposed adaptive burst mode control, there are three basic operation modes: Pcell mode (PCM), naturalbalancemode (NBM), and Ncell mode (NCM). The basic operational waveforms are shown in
Fig. 8
.
Simplified control block diagram of burst mode control based on the bidirectional dual buckboost voltage balancer.
The basic operational waveforms of three operation modes in the bidirectional dual buckboost voltage balancer.
 D. Three Operation Modes of the Proposed Burst Mode Control
1) Pcell mode (PCM):
When
R_{1}
is bigger than
R_{2}
,
V_{dc1}
increases and
V_{dc2}
decreases while
V_{dc}
is kept constant at the same time. After
V_{dc2}
touches the voltage lower limit
V_{lower}
, the Pcell leg switch
S
_{1}
works under the current mode control with the preset
i_{Lref}
. Then,
V_{dc2}
begins to increase to the allowed voltage lower limit
V
_{lowerallowed}
during a burstmode switching period, in which the voltage difference should satisfy the condition of
V_{upperallowed}

V
_{lowerallowed}
˂
V_{dc1}

V_{dc2}
˂
V_{upper}

V_{lower}
. After that,
S
_{1}
does not work until
V
_{dc2}
reaches
V_{lower}
again. Operating in the Pcell mode, the positive inductor current
i_{L}
transfers the redundant power from the upside to the downside, as shown in
Fig. 8
(a).
2) Naturalbalance mode (NBM):
As shown in
Fig. 8
(b), when
V_{dc1}
and
V_{dc2}
are naturally maintained between the allowed voltage lower limit
V_{lowerallowed}
and upper limit
V_{upperallowed}
without control, the voltage balancer does not work and
i_{L}
is zero at this time.
3) Ncell mode (NCM):
When
R_{1}
is smaller than
R_{2}
,
V_{dc1}
decreases and
V_{dc2}
increases while
V_{dc}
is kept constant at the same time. After
V_{dc2}
touches the voltage upper limit
V_{upper}
, the Ncell leg switch
S_{2}
works under the current mode control with the preset
i_{Lref}
. Then,
V_{dc2}
begins to decrease to the allowed voltage upper limit
V_{upperallowed}
during a burst mode switching period, in which the voltage difference should satisfy the condition of
V_{upperallowed}

V_{lowerallowed}
˂
V_{dc2}

V_{dc1}
˂
V_{upper}

V_{lower}
After that,
S_{2}
does not work until
V_{dc2}
reaches
V_{upper}
again. Operating in the Ncell mode, the negative
i_{L}
transfers redundant power from the downside to the upside as shown in
Fig. 8
(c).
Because
V_{dc1}
and
V_{dc2}
should not higher than
V_{upper}
or less than
V_{lower}
, the switches (
S
_{1}
,
S
_{2}
) will work only when the dc voltage values (
V_{dc1}
,
V_{dc2}
) do not meet predetermined values. Then, a positive or negative intermittent
i_{L}
will exist to transfer the corresponding power between the two poles making the voltages balanced. Thus, the switching loss of the balancer will be effectively reduced and its efficiency will be improved. In addition, the switches of the Pcell leg and Ncell leg are controlled separately. This can avoid the power switches shootthrough problem in traditional bridge topological structures. In this case, high frequency MOSFETs switches can be used to reduce the system volume.
 E. Realization of the Proposed Burst Mode Control Scheme
The realization of the proposed burst mode control scheme is presented in
Fig. 9
.
V_{ref}
is equal to half of the total dc voltage value
V_{dc}
. If
R_{1}
is bigger than
R_{2}
,
V
_{dc1}
increases combined with the decrease of
V_{dc2}
. When
V
_{dc2}
is less than
V_{lower}
, the following is true
V_{ref}

V_{dc2}
>
V_{ref}

V_{lower}
and
V_{ref}

V_{dc2}
>
V_{ref}

V
_{lowerallowed}
. At this time, the outputs of the two comparators in the Pcell mode are 1 and 0, respectively, and the outputs of the flipflop and the OR gate are 1. The PWM current control regulates
i_{L}
tracking the reference current
i_{Lref}
, with the outputs of the comparator and the OR gate sent to the AND gate together. When the output of the AND gate connected with the comparator is 1, a trigger signal will be sent to switch
S
_{1}
which will be turned on with a positive
i_{L}
. Operating in the Pcell mode, the Pcell leg will transfer redundant power from the upside to the downside, and then
V
_{dc2}
will increase. When
V_{dc2}
goes up to the value of
V
_{lowerallowed}
, there will be
V_{ref}

V_{dc2}
<
V_{ref}

V_{lower}
and
V_{ref}

V_{dc2}
˂
V_{ref}

V_{lowerallowed}
. At this time, the outputs of the two comparators are 0 and 1, respectively. The output of the flipflop is zero and the trigger signal of
S
_{1}
disappears. Then,
S
_{1}
is turned off and the voltage balancer stops working. As the operating procedures of
S
_{2}
are the same as those of
S
_{1}
, the analysis process of
S
_{2}
is not given out.
Realization of the proposed burst mode control scheme.
 F. Main Current Relationships of the Bidirectional Dual BuckBoost Voltage Balancer
If
R_{1}
is bigger than
R_{2}
,
V_{dc1}
goes up and
V_{dc2}
decreases at the same time. Since
V_{dc}
is kept constant through regulation by the bidirectional rectifier, the ripple voltage Δ
V_{dc1}
of
V_{dc1}
is also equal to the negative ripple voltage Δ
V_{dc2}
of
V_{dc2}
.
If the two capacitors,
C_{u}
and
C_{d}
, are equal,
i_{Cu}
=
i_{Cd}
.
Since
i_{L}
=
i_{Cd}
+
i
_{R}
_{2}

i_{Cu}

i
_{R}
_{1}
, the following can be obtained:
If (
i_{L}
+
V_{dc1}
/
R
_{1}
) ×
R
_{2}
>
V_{ref}
is satisfied, either the voltage balancer will work normally to balance
V_{dc1}
and
V_{dc2}
, or the balancer will fail to work properly to balance
V_{dc1}
and
V_{dc2}
. On the premise of the balancer working normally, the balancer does not work all the time if (
i_{L}
+
V_{upperallowed}
/
R
_{1}
) ×
R
_{2}
>
V_{lowerallowed}
is satisfied, and an intermittent inductor current
i_{L}
will exist. However, the balancer will always at work with a sustained
i_{L}
if (
i_{L}
+
V_{upperallowed}
/
R
_{1}
) ×
R
_{2}
=
V_{lowerallowed}
is satisfied. When
R
_{1}
is infinity in extreme cases,
R
_{2}
should not be less than
V_{ref}
/
i_{Lref}
, or the balancer will not work.
IV. THE AVERAGE SMALLSIGNAL MODEL FOR THE BIDIRECTIONAL DUAL BUCKBOOST VOLTAGE BALANCER
In order to select the control system parameters, an average smallsignal model of the voltage balancer under CCM is derived. The duty cycles of
S
_{1}
and
S
_{2}
are defined as
, respectively, where
D
_{1}
,
D
_{2}
,
are the stable duty ratios and the perturbations of
d
_{1}
and
d
_{2}
. Moreover, the voltage
v_{dc2}
and the inductance current
i_{L1}
are defined as
are the stable voltage values, the current values and the perturbations of
v_{dc2}
and
i_{L1}
.
From
Fig. 6
, it is possible to obtain (5) and (6) when the Pcell bridge leg operates under CCM.
(1)
S
_{1}
turning on
(2)
S
_{1}
turning off
According to the methods of building the average model, it can be derived from (5) and (6):
Thus, the transfer function of the inductance current
i_{L1}
versus the duty cycle
d
_{1}
is represented by:
From
Fig. 8
, (9) and (10) can be obtained when the Ncell bridge leg operates under CCM.
(1)
S
_{2}
turning on
(2)
S
_{2}
turning off
According to the methods of building the average model, (9) and (10) are rewritten by:
Thus, the transfer function of the inductance current
i_{L2}
versus the duty cycle
d
_{2}
is represented by:
Comparing (8) with (12), the openloop transfer functions of the Pcell and Ncell bridge legs are the same. Using the following equation
G
_{2}
(
s
) =
G
_{1}
(
s
)/[1 +
G
_{1}
(
s
)], the closedloop transfer functions
G_{2}
(
s
) can also be achieved. The control system diagram of the voltage balancer can be illustrated by
Fig. 10
.
Control system based on the current mode control.
V. SIMULATION RESULTS FOR THE BIDIRECTIONAL DUAL BUCKBOOST VOLTAGE BALANCER
To validate the aforementioned analysis, computer simulations of the output voltages and inductance currents are carried out in PISM software. The total power of the voltage balancer is 10kW and the switching frequency is 30 kHz. The other main simulation parameters are listed as follows:
V_{dc}
= 400V,
V_{ref}
= 200V,
L
_{1}
=
L
_{2}
= 0.2mH,
i_{Lref}
= 50A,
C_{u}
=
C_{d}
= 10mF,
V_{upper}
= 202.2V,
V_{lower}
= 197.8V,
V_{upperallowed}
= 201.8V, and
V
_{lowerallowed}
= 198.2V.
R_{dc1}
and
R_{dc2}
will be given according to the simulation requirements in the following.
 G. Analysis of the Control System Parameters
The PI parameters of the control system in
Fig. 10
are
K_{p}
= 3.7 and
K_{i}
= 0.001. With the above simulation parameters, a Bode diagram of the control system is shown in
Fig.1 1
, where
G
_{1}
and
G
_{2}
represent the openloop and closedloop transfer functions, respectively. When the unbalanced loads are
R
_{1}
= 5Ω and
R
_{2}
= 50MΩ, that is to say,
R
_{2}
is infinite relative to
R
_{1}
, the openloop transfer function
G
_{1}
(
s
) and the closedloop transfer function
G
_{2}
(
s
) are described in (13) and (14), respectively. In this case, a Bode diagram of the control system is shown in
Fig. 11
(a). When the unbalanced loads are
R
_{1}
= 500Ω and
R
_{2}
= 50MΩ, the openloop transfer function
G
_{1}
(
s
) and the closedloop transfer function
G
_{2}
(
s
) are described in (15) and (16), respectively. In this case, a Bode diagram of the control system is shown in
Fig. 11
(b). By analyzing the amplitudefrequency characteristic and phasefrequency response curves in the Bode diagram, where G
_{m}
= inf and P
_{m}
= inf indicate that the gain margin and phase margin are infinite, it can be seen that the given control system is stable.
Bode diagram of the openloop and closedloop system transfer function: (a) R_{1} = 5Ω, R_{2} = 50MΩ; (b) R_{1} = 500Ω, R_{2} = 50MΩ.
 H. Simulation Results
The simulation waveforms of the output voltages (
V_{dc1}
,
V_{dc2}
) and inductor current
i
_{L}
are given in
Fig. 12
. As in the above analysis, the simulations are carried out in three different operation modes.
Simulation results of the voltage and current relationships: (a) Pcell mode; (b) Naturalbalance mode; (c) Ncell mode.
Pcell mode: When
R_{1}
is bigger than
R
_{2}
(
R_{1}
= 50MΩ,
R_{2}
= 5Ω),
V_{dc1}
will be higher than
V_{dc2}
. The Pcell leg switch
S
_{1}
works under the current control mode with a preset
i_{Lref}
of 50A and it generates an intermittent positive current
i_{L}
to transfer the redundant power from the upside to the downside. With a decrease of
V_{dc1}
and an increase of
V_{dc2}
, both
V_{dc1}
and
V_{dc2}
come back to allowable values during a burst mode switching period. Then, the voltage balancer stops working and the inductor current
i_{L}
disappears. Waveforms of the voltages and inductor current are shown in
Fig. 12
(a). To get a clear view of the voltages and current,
Fig. 12
(a) is partially enlarged, as shown in
Fig. 13
.
Detailed simulation results of the voltage and current relationships in Pcell mode.
Naturalbalance mode: When
R
_{1}
is equal to
R_{2}
(
R_{1}
=
R_{2}
= 5Ω),
V_{dc1}
and
V_{dc2}
automatically tend to become balanced without control, and ultimately they are kept equal as in
Fig. 12
(b). Because
V_{dc1}
and
V_{dc2}
naturally keep between
V_{lowerallowed}
and
V_{upperallowed}
, the balancer does not work and the inductor current maintains a constant of zero in this case.
Ncell mode: When
R_{1}
is smaller than
R_{2}
(
R_{1}
= 5Ω,
R_{2}
= 50MΩ),
V_{dc1}
is lower than
V_{dc2}
. Thus, the curve of
V_{dc1}
has a downward trend when comparing with
Fig. 12
(a) and
Fig. 12
(b). Similarly, the Ncell leg switch
S
_{2}
works under the current control mode with a preset
i_{Lref}
of 50A and it generates a negative intermittent current
i_{L}
to transfer redundant power from the downside to the upside. With an increase of
V_{dc1}
and a decrease of
V_{dc2}
, both
V_{dc1}
and
V_{dc2}
come back to allowable values, and the voltage balancer stops working and the inductor current disappears. The waveforms of the voltages and current are shown in
Fig. 12
(c).
VI. EXPERIMENTAL RESULTS
To demonstrate the viability of the proposed voltage balancer and the novel burstmode control scheme, a 5kW experiment system, shown in
Fig. 14
, was designed and tested. The whole system includes a 400V ac input, voltage balancer, dualoutputs full bridge inverter and loads which can be divided into dc loads and ac loads. Additionally, the ac loads include an ac resistive load and a halfbridgerectifier ac load (such as a residential heater) which will lead to an unbalance between
V_{dc1}
and
V_{dc2}
. Thus, the experiment consists of two parts: the first part is a dc load test which is emulated by two pure resistors; and the second part is an ac load test including the balanced resistive load and unbalanced load of the halfbridge resistive heater. The simulation parameters are listed as follows:
C_{u}
=
C_{d}
= 10mF,
L
_{1}
=
L
_{2}
= 0.2mH,
C_{1}
=
C_{2}
= 40uF,
R_{ac1}
=
R_{ac2}
= 25Ω, and
i_{Lref}
= 30A. A directly coupled inductor is used in the voltage balancer to reduce its volume and weight, and
L_{d1}
=
L_{d2}
= 50uH,
L_{com}
= 150uH. The resistance of the halfbridgerectifier load is 11Ω.
R_{dc1}
and
R_{dc2}
will be given according to the experimental requirements in the following.
The experiment system structure.
 I. Experimental Results Analysis of the dc Load
The first experiment is conducted with unbalanced dc loads so that
R_{dc1}
= ∞ and
R_{dc2}
= 9Ω, 18Ω. The steadystate experimental waveforms of the inductor current
i_{L}
and load current
i_{dc2}
, which is the current through
R_{dc2}
, are shown in
Fig. 15
. As shown in
Fig. 15
, the switch duty cycle is reduced with the increase of
R_{dc2}
, and the duration of
i_{L}
becomes shorter. Even in this extreme condition, the novel burstmode control scheme and voltage balancer work well to provide an appropriate voltage for
R_{dc2}
.
The experimental waveforms of the inductor current and load current. (a) R_{dc1} = ∞, R_{dc2} = 9Ω; (b) R_{dc1} = ∞, R_{dc2} = 18Ω.
 J. Experimental Results Analysis of the ac Load
The second experiment is conducted with unbalanced ac loads which are simulated by a halfbridgerectifier load and a resistive load. For security reasons, a small
V_{dc}
is applied to observe the system performance when the voltage balancer is first disconnected.
Fig. 16
shows the experimental waveforms of the ac voltage
v_{out2}
and the halfbridgerectifier load current
i_{out1}
. It can be seen that the halfbridgerectifier load leads to a serious asymmetric output ac voltage
v_{out2}
without the balancer.
The experimental waveforms without the voltage balancer when connected to unbalanced ac load.
Fig. 17
shows the steadystate output ac voltages (
v_{out1}
,
v_{out2}
), and the halfbridge heater current
i_{out1}
when the voltage balancer is connected. It can be seen that the dualoutput voltages have almost the same RMS values,
v_{out1}
= 118.90V and
v_{out2}
= 117.52V, though the current of the halfbridge heater is obviously unsymmetrical.
The experimental waveforms of the voltage and current with the voltage balancer when connected the unbalanced ac load.
Fig. 18
and
19
show the experimental transition waveforms of the dualoutput ac voltages, and the load currents between the balanced and unbalanced loads. No impact voltages or currents appear in the system with a smooth transition, and the output voltages are balanced and almost the same before and after the loads change.
The experimental transition waveforms of the ac voltage and current from balanced load to unbalanced load.
The experimental transition waveforms of the ac voltage and current from unbalanced load to balanced load.
Fig. 20
and
21
show the experimental waveforms of the dualoutputs ac voltages (
v_{out1}
,
v_{out2}
), the inductor current
i_{L}
and the halfbridge heater current
i_{out1}
. The inductor current is zero when the ac loads are balanced. However, it becomes an intermittent negative current to balance the two voltages when the ac loads are unbalanced. Similarly, there are no impact voltages or currents in the system, and the output voltages are kept balanced before and after the loads change with a smooth transition.
The experimental waveforms of voltage and current with the voltage balancer from balanced load to unbalanced load.
The experimental waveforms with voltage balancer from unbalanced load to balanced load.
According to the above experimental results, the output ac voltages (
v_{out1}
,
v_{out2}
) have almost the same values with the control of the voltage balancer, whether balanced loads or unbalanced loads are connected. Thus, the bidirectional dual buckboost voltage balancer with the burstmode control scheme is shown to be correct.
VII. CONCLUSION
In this paper, a bidirectional dual buckboost voltage balancer and a burstmode control scheme are proposed for lowvoltage bipolartype DC microgrids. The structure of the voltage balancer will enhance the system reliability thanks to the fact that it does not have any shootthrough problems. Under the proposed novel burstmode control scheme, the balancer works and generates an intermittent inductor current with lower power losses to balance the two output DC voltages when the two voltages do not satisfy predetermined values. At last, simulation and experimental results have verified that the bidirectional dual buckboost voltage balancer based on the burstmode control scheme has a good ability to balance the output voltages with balanced or unbalanced DC/AC loads. Moreover, the system has a smooth transition with no impact voltages or currents when the loads change.
BIO
Chuang Liu received his M.S. degree in Electrical Engineering from Northeast Dianli University, Jilin, China, in 2009, and his Ph.D. degree from the Harbin Institute of Technology, Harbin, China, in 2013. From 2010 to 2012, he was with the Future Energy Electronics Center (FEEC), Virginia Tech, Blacksburg, VA, USA, as a Visiting PhD Student, supported by the Chinese Scholarship Council. Since 2013, he has been an Associate Professor in the School of Electrical Engineering of Northeast Dianli University, Jilin, China. His current research interests include intelligent universal transformers (IUT) for renewable energy systems, bus power flow controllers (BPFC) for hybrid AC/DC distribution, PHEV/PEV smart parking lot/building charging systems, battery energy storage systems and wireless power transfer.
Dawei Zhu received his B.S. degree from Northeast Dianli University, Jilin, China, in 2013, where he is presently working towards his M.S. degree in the Department of Electrical Engineering. His current research interests include distributed generation integration and smart power grids.
Jia Zhang received her B.S. and M.S. degrees, from Northeast Dianli University, Jilin, China, in 2012 and 2015, respectively. She is presently working for the State Grid, Jilin Electric Power Training Center, Jilin, China. Her current research interests include smart power grids and power electronics.
Haiyang Liu received his B.S. degree from the Huazhong University of Science and Technology, Wuhan, China, in 2012. He is presently working towards his M.S. degree in the Department of Electrical Engineering, Northeast Dianli University, Jilin, China. His current research interests include intelligent universal transformers (IUT) for renewable energy systems.
Guowei Cai received his B.S. and M.S. degrees from Northeast Dianli University, Jilin, China, in 1990 and 1993, respectively, and his Ph.D. degree from the Harbin Institute of Technology, Harbin, China, in 1999. Since 2004, he has been a Professor in the School of Electrical Engineering of Northeast Dianli University. His current research interests include power system transient stability analysis and electrical power markets.
Kakigano H.
,
Miura Y.
,
Ise T.
2010
“Lowvoltage bipolartype dc microgrid for super high quality distribution,”
IEEE Trans. Power Electron.
25
(12)
3066 
3075
DOI : 10.1109/TPEL.2010.2077682
Lago J.
,
Heldwein M. L.
2011
“Operation and controloriented modeling of a power converter for current balancing and stability improvement of DC active distribution networks,”
IEEE Trans. Power Electron.
26
(3)
877 
885
DOI : 10.1109/TPEL.2011.2105284
Kanchev H.
,
Lu D.
,
Colas F.
,
Lazarov V.
,
Francois B.
2011
“Energy management and operational planning of a microgrid with a PVbased active generator for smart grid applications,”
IEEE Trans. Ind. Electron.
58
(10)
4583 
4592
DOI : 10.1109/TIE.2011.2119451
Byeon G.
,
Yoon T.
,
Oh S.
,
Jang G.
2013
“Energy management strategy of the DC distribution system in buildings using the EV service model,”
IEEE Trans. Power Electron.
28
(4)
1544 
1554
DOI : 10.1109/TPEL.2012.2210911
Seo G.S.
,
Lee K.C.
,
Cho B.H.
2013
"A new DC antiislanding technique of electrolytic capacitorless photovoltaic interface in DC distribution systems,”
IEEE Trans. Power Electron.
28
(4)
1632 
1641
DOI : 10.1109/TPEL.2012.2208226
Leu C. S.
,
Nha Q. T.
2013
“A halfbridge converter with input current ripple reduction for DC distribution systems,”
IEEE Trans. Power Electron.
28
(4)
1756 
1763
DOI : 10.1109/TPEL.2012.2213269
Wu T. F.
,
Chang C. H.
,
Lin L. C.
,
Yu G. R.
,
Chang Y. R.
2013
“DCbus voltage control with a threephase bidirectional inverter for DC distribution systems,”
IEEE Trans. Power Electron.
28
(4)
1890 
1899
DOI : 10.1109/TPEL.2012.2206057
Anand S.
,
Fernandes B. G.
,
Guerrero M.
2013
“Distributed control to ensure proportional load sharing and improve voltage regulation in lowvoltage DC microgrids,”
IEEE Trans. Power Electron.
28
(4)
1900 
1913
DOI : 10.1109/TPEL.2012.2215055
Loh P. C.
,
Li D.
,
Chai Y. K.
,
Blaabjerg F.
2013
“Autonomous operation of hybrid microgrid with AC and DC subgrids,”
IEEE Trans. Power Electron.
28
(5)
2214 
2223
DOI : 10.1109/TPEL.2012.2214792
Kakigano H.
,
Miura Y.
,
Ise T.
2013
“distribution voltage control for DC microgrids using fuzzy control and gainscheduling technique,”
IEEE Trans. Power Electron.
28
(5)
2246 
2258
DOI : 10.1109/TPEL.2012.2217353
Anand S.
,
Fernandes B. G.
2013
“Reducedorder model and stability analysis of lowvoltage DC microgrid,”
IEEE Trans. Ind. Electron.
60
(11)
5040 
5049
DOI : 10.1109/TIE.2012.2227902
Grillo S.
,
Musolino V.
,
Piegari L.
,
Tironi E.
,
Tornelli C.
2014
“DC islands in AC smart grids,”
IEEE Trans. Power Electron.
29
(1)
89 
98
DOI : 10.1109/TPEL.2013.2251666
Dragicevic T.
,
Guerrero J.M.
,
Vasquez J.C.
,
Skrlec D.
2014
“Supervisory control of an adaptivedroop regulated DC microgrid with battery management capability,”
IEEE Trans. Power Electron.
29
(2)
695 
706
DOI : 10.1109/TPEL.2013.2257857
Chen L. H.
,
Peng F. Z.
2008
“Deadtime elimination for voltage source inverters,”
IEEE Trans. Power Electron.
23
(2)
574 
580
DOI : 10.1109/TPEL.2007.915766
Li S. N.
,
Tolbert L. M.
,
Wang F.
,
Peng F. Z.
“Reduction of stray inductance in power electronic modules using basic switching cells,”
Energy Conversion Congress and Exposition (ECCE)
2010
2686 
2691
Li S. G.
,
Tolbert L. M.
,
Wang F.
,
Peng F. Z.
“Pcell and Ncell based IGBT module: Layout design, parasitic extraction, and experimental verification,”
in IEEE Applied Power Electronics Conference and Exposition (APEC)
2011
372 
378
Floricau D.
,
Floricau E.
,
Gateau G.
2011
“New multilevel converters with coupled inductors: properties and control,”
IEEE Trans. Ind. Electron.
58
(12)
5344 
5351
DOI : 10.1109/TIE.2011.2112319
Yao Z.
,
Xiao L.
,
Yan Y.
2009
“Control strategy for series and parallel output dualbuck half bridge inverters based on DSP control,”
IEEE Trans. Power Electron.
24
(2)
434 
444
DOI : 10.1109/TPEL.2008.2007117
Yao Z.
,
Xiao L.
,
Yan Y.
2009
“Dualbuck fullbridge inverter with hysteresis current control,”
IEEE Trans. Ind. Electron.
56
(8)
3153 
3160
DOI : 10.1109/TIE.2009.2022072
Sun P. W.
,
Liu C.
,
Lai J.S.
,
Chen C.L.
2012
“Cascade dual buck inverter with phaseshift control,”
IEEE Trans. Power Electron.
27
(4)
2067 
2077
DOI : 10.1109/TPEL.2011.2169282
Sun P.
,
Liu C.
,
Lai J.S.
,
Chen C.L.
2012
“Gridtie control of cascade dualbuck inverter with widerange power flow capability for renewable energy applications,”
IEEE Trans. Power Electron.
27
(4)
1839 
1849
DOI : 10.1109/TPEL.2011.2175009
Sun P.
,
Liu C.
,
Lai J.
,
Chen C.
,
Kees N.
2012
“Threephase dualbuck inverter with unified pulsewidth modulation,”
IEEE Trans. Power Electron.
27
(3)
1159 
1167
DOI : 10.1109/TPEL.2011.2164269
Liu C.
,
Sun P.
,
Lai J.S.
,
Ji Y.
,
Wang M.
,
Chen C.L.
,
Cai G.
2012
“Cascade dualboost/buck activefrontend converter for intelligent universal transformer,”
IEEE Trans. Ind. Electron.
59
(12)
4671 
4680
DOI : 10.1109/TIE.2011.2182014
Zhang X.
,
Yao C.
,
Guo F.
,
Wang J.
“Optimal operation and burstmode control for improving the efficiency of the quasiswitchedcapacitor resonant converter,”
Energy Conversion Congress and Exposition (ECCE)
2014
5444 
5450
Vasic D.
,
Liu Y.P.
,
Costa F.
,
Schwander D.
“Piezoelectric transformerbased DC/DC converter with improved burstmode control,”
Energy Conversion Congress and Exposition (ECCE)
2013
140 
146