new DC-DC power converter is researched for renewable energy and battery hybrid power supplies systems in this paper. At the charging mode, a renewable energy source provides energy to charge a battery via the proposed converter. The operating principle of the proposed converter is the same as the conventional DC-DC buck converter. At the discharging mode, the battery releases its energy to the DC bus via the proposed converter. The proposed converter is a non-isolated high step-up DC-DC converter. The coupled-inductor technique is used to achieve a high step-up voltage gain by adjusting the turns ratio. Moreover, the leakage-inductor energies of the primary and secondary windings can be recycled. Thus, the conversion efficiency can be improved. Therefore, only one power converter is utilized at the charging or discharging modes. Finally, a prototype circuit is implemented to verify the performance of the proposed converter. The use of the fossil fuels causes environmental pollution and ecological problems. In addition, carbon-dioxide emissions result in global warming. Therefore, the development and application of renewable energy sources has become very important [1] - [10] . Renewable energies include wind power, solar power, ocean energy, hydrogen energy, hydro energy and so on. Fig. 1 (a) shows the conventional renewable energy and battery hybrid power supply system. It can be seen that the battery is charged from a renewable energy source via two converters [11] - [17] . Therefore, the conventional hybrid power supply system results in energy waste. For this reason, this paper presents a new DC-DC converter for three-port renewable energy and battery hybrid power supplies systems, as shown in Fig. 1 (b). It can be seen that the battery is only via one converter at the charging or discharging mode. Fig. 2 shows the circuit configuration of the proposed converter. The pulse-width modulation (PWM) technique is used for the proposed converter. At the charging mode, the renewable energy source provides its energy to charge the battery via the proposed converter. Here the proposed converter is a conventional DC-DC buck converter [18] , [19] . At the discharging mode, the battery releases its energy to the DC bus via the proposed converter. Meanwhile, this converter is a non-isolated high step-up DC-DC converter [20] . The coupled-inductor technique is used to achieve a high step-up voltage gain by adjusting the turns ratio. Moreover, the leakage-inductor energies of the primary and secondary windings can be recycled. Thus, only one power converter is utilized at the charging and discharging modes. The operating principles and steady-state analyses will be described in the following sections for the charging and discharging modes. In order to analyze the steady-state characteristics of the proposed converter, some conditions are assumed: (1) Because the capacitors C ₁ , C_B , and C_bus are sufficiently large, the voltages across these capacitors can be treated as constant during each switching period. (2) The ON-state resistance of the switches, the forward voltage drop of the diodes, and the equivalent series resistances of the coupled inductor and capacitors are ignored. When the proposed converter is operated in the charging mode, the primary winding of the coupled inductor is used for a general inductor. The equivalent circuit of the proposed converter at the charging mode is shown in Fig. 3 , where R_B is the equivalent load for the battery. The PWM technique is used to control switch S ₁ . Switch S ₂ is used for the synchronous rectifier. Fig. 4 shows typical waveforms of the proposed converter with the continuous conduction mode (CCM) operation at the charging mode. The operating principles and steady-state analysis are described as follows: 1) Mode 1: During this time interval [ t ₀ , t ₁ ], switch S ₁ is turned on and switch S ₂ is turned off. The current flow path is shown in Fig. 5 (a). The renewable-energy source V_in supplies its energy to for the coupled inductor L_m , capacitor C_B , and load R_B . Meantime, the coupled inductor L_m stores its energy. Thus, the current i_Lm is increased. The voltage across the coupled inductor L_m is given by: 2) Mode 2: During this time interval [t ₁ , t ₂ ], switch S ₁ is turned off and switch S ₂ is turned on. The current flow path is shown in Fig. 5 (b). Meanwhile, switch S ₂ is utilized for a synchronous rectifier. The energy stored in the coupled inductor L _m is released to the capacitor C _B and load R _B . The voltage across the coupled inductor L _m is derived as: By using the voltage-second balance principle on the coupled inductor L_m , the following equation is obtained: Substituting (1) and (2) into (3), the voltage gain in the charging mode is found to be: When the proposed converter is operated in the boundary conduction mode (BCM), typical waveforms are shown in Fig. 6 . Thus, the peak value of the coupled-inductor current is written as: By utilizing the ampere-second balance principle on the capacitor C_B , the following equation can be expressed as: From (6), the peak value of the coupled-inductor current can be rewritten as: Then, the normalized inductor time constant is defined as: Substituting (4), (5), and (8) into (7), the boundary normalized inductor time constant is given by: If τ _Lm1 is larger than τ _Lm1,B , the proposed converter is operated in the CCM at the charging mode. The equivalent circuit of the proposed converter at the discharging mode is shown in Fig. 7 , where R_L is the equivalent load for the DC bus. The PWM technique is used to control switch S ₂ . Switch S ₁ is turned off at the discharging mode. Fig. 8 shows typical waveforms of the proposed converter with the CCM operation at the discharging mode. The operating principles and steady-state analysis are described as follows: 1) Mode 1: During this time interval [ t ₀ , t ₁ ], switch S ₂ is turned on. The current flow path is shown in Fig. 9 (a). The energy of the battery is released to the magnetizing inductor L_m and primary leakage inductor L _k1 . Thus, the currents i_Lm and i _Lk1 are increased. The secondary leakage inductor L _k2 , secondary winding N ₂ , capacitor C ₁ , and battery are in series to release their energies for the load R_L . The energy of the capacitor C_bus is also provided to the load R_L . Therefore, the current i _Lk2 decreases. At i _Lk2 = 0, the energy stored in the leakage inductor L _k2 is completely recycled to the load R_L . 2) Mode 2: During this time interval [t ₁ , t ₂ ], switch S ₂ is still turned on. The current flow path is shown in Fig. 9 (b). The energy of the battery is still released to the magnetizing inductor L _m and primary leakage inductor L _k1 . The energy of the capacitor C _bus is provided to the load R _L . 3) Mode 3: During this time interval [t ₂ , t ₃ ], switch S ₂ is turned off. The current flow path is shown in Fig. 9 (c). The energies stored in the magnetizing inductor L _m and primary leakage inductor L _k1 are released the capacitor C ₁ . Thus, the currents i _Lm and i _Lk1 are decreased. The secondary winding N ₂ , capacitor C ₁ , and battery are in series to release their energies for the leakage inductor L _k2 , capacitor C _bus , and load R _L . Therefore, the current i _Lk2 is increased. At i _Lk1 = 0, the energy stored in the leakage inductor L _k1 is completely recycled to the capacitor C ₁ . 4) Mode 4: During this time interval [t ₃ , t ₄ ], switch S ₂ is still turned off. The current flow path is shown in Fig. 9 (d). The secondary leakage inductor L _k2 , secondary winding N ₂ , capacitor C ₁ , and battery are in series to release their energies for the capacitor C _bus and load R _L . Therefore, the currents i _Lm and i _Lk2 are decreased. When switch S ₂ is turned on, the following equation can be represented as: Thus: where the turns ratio of the coupled inductor n = N ₂ / N ₁ , and the coupled coefficient k = L_m /( L_m + L _k1 ). From the operating principle, it is known that the energy stored in the primary leakage inductor L _k1 is recycled to the capacitor C ₁ . By using the ampere-second balance principle on the capacitor C ₁ , the released time duration of the primary leakage-inductor energy can be expressed as [20] : By utilizing the voltage-second balance principle on the primary leakage inductor L _k1 and magnetizing inductor L_m , the following equations can be obtained: where v _Lk1(tr) is the voltage across the primary leakage inductor L _k1 during the time duration [ t ₂ , t ₃ ] and the voltage v _N1(OFF) across the magnetizing inductor L_m during the time duration [ t ₂ , t ₄ ]. Substituting (11) and (14) into (15), yields: By substituting (12) into (16), it is possible to derive: From Fig. 9 (c), the following equation can be obtained: Substituting (17) and (18) into (19), the voltage across the capacitor C ₁ is written as follows: During the switch S ₂ OFF-period, the following voltage equation is found as: Therefore: Using the voltage-second balance principle on the secondary winding of the coupled inductor N ₂ , the equation can be represented as: Substituting (13), (20), and (22) into (23), the voltage gain can be found as follows: At k = 1, equation (24) is rewritten as: The voltage gain with parasitic components is analyzed as follows. In order to simplify the analysis, the leakage inductors of the coupled inductor are neglected. The equivalent circuit is shown in Fig. 10 . r _N1 and r _N2 represent the equivalent series resistances (ESR) of the primary and secondary windings of the coupled inductor. V _FD2 and r _D2 are the ON-state forward voltage drop and resistance of D ₂ . r _S2 denotes the ON-state resistance of S ₂ . When switch S ₂ is turned on, the equivalent circuit is shown in Fig. 10 (a). The average values of i_bus and v _N1 are written as: When switch S ₂ is turned off, the equivalent circuit is shown in Fig. 10 (b). The average values of i_bus and v _N1 are found as: Since the leakage inductors of the coupled inductor are neglected, the coupling coefficient k is equal to 1. From (20), the voltage V _c1 can be rewritten as: Substituting (30) into (29), yields: By using the ampere-second balance principle on C_bus , the following equations are obtained as: Substituting (26) an (28) into (32), I _b(OFF) is derived as: In addition, I _b(ON) can be given as: Using the volt-second balance principle on L_m , yields: Substituting (27), (31), (33), and (34) into (35), the actual voltage gain can be obtained as follows: The curves of the ideal and actual voltage gain under n =3, r _N1 = r _N2 =100 mΩ, r _D2 =50 mΩ, r _S2 =50 mΩ, V _FD2 =1.25 V, V_bat =24 V, and R_L =200 Ω are plotted in Fig. 11 . It can be seen that the proposed converter in the discharging mode can achieve a high step-up voltage gain. When the proposed converter is operating in the BCM, typical waveforms are shown in Fig. 12 . Thus, the peak value of the magnetizing-inductor current is given as: Applying the ampere-second balance principle on the capacitor C_bus , the following equation is found: From the above equation, the peak value of the magnetizing-inductor current is rewritten as: Then, the normalized inductor time constant is defined as: Substituting (24), (37), (40) into (39), the boundary inductor time constant is obtained as follows: At k = 1, τ _Lm2 can be rewritten as: If τ _Lm2 is larger than τ _Lm2,B , the proposed converter in the discharging mode is operated in the CCM. In order to verify the feasibility of the proposed converter, a prototype circuit is built for a fuel-cell and battery hybrid power supply system. The electric specifications and circuit components are selected as the input voltage V_in = 28~36.5 V, output power P_o = 200~20 W, battery voltage V_bat = 24 V, DC-bus voltage V_bus = 200 V, switching frequency f_s = 50 kHz, coupled inductor L_m = 72 μ H and n = 3, capacitors C_B = C ₁ = C_bus = 220 μ F, switches S ₁ (IXFH80N085) and S ₂ (IXTQ96N20P), and diodes D ₁ (DSSK60-02AR) and D ₂ (DSEP30-06A). Fig. 13 shows the experimental results at the charging mode. The electrical specifications are V_in = 28 V, V_bat = 24 V, and full load P_o = 200 W. From Figs. 13 (a) and 13 (c), it can be seen that the current waveforms, i _S1 and i_Lm , are same during the S ₁ ON-period. As can be seen from Figs. 13 (b) and 13 (c), the current waveforms, i _S2 and i_Lm , are the same during the S ₁ OFF-period. Fig. 13 (c) shows that the proposed converter is operated in the CCM. The measured efficiency at the charging mode is shown in Fig. 15 . The measured efficiency is around 95.2%~97.5% under V_in = 28~36.5 V and P_o = 200~20 W. Fig. 14 shows the experimental results at the discharging mode. The electrical specifications are V_bat = 24 V, V_bus = 200 V, and full load P_o = 200 W. The waveforms, v _gs2 , v _S2 , and i _S2 , are shown in Fig. 14 (a). It can be seen from the waveform i _S2 that the proposed converter is operated in the CCM. From the waveform v _S2 , the voltage across S ₂ is clamped at approximately 70 V during the S ₂ OFF-period. Therefore, a low rated voltage MOSFET can be adopted to reduce the conduction loss. As shown in Fig. 14 (b), the voltage across D ₁ is clamped at approximately 70 V during the S ₂ OFF-period. The waveforms, i _D1 and i _Lk1 , are shown in Figs. 14 (b) and 14 (c). It can be seen that they consist with the operating principle. The measured efficiency at the discharging mode is shown in Fig. 15 . The measured efficiency is around 93.3%~95.4%. In the conventional hybrid power supplied system, two power converters are used for the battery charging or discharging modes. The conventional system results in energy waste. A new DC-DC converter for renewable energy and battery hybrid power supplied systems is investigated in this paper. At the charging mode, a renewable energy source can provide its energy to charge the battery via the proposed converter. At the discharging mode, the battery can release its energy to the DC bus via the proposed converter. Thus, only one power converter is utilized at the charging or discharging modes. The proposed converter can increase the conversion efficiency. The measured efficiency is around 95.2%~97.5% at the charging mode and it is around 93.3%~95.4% at the discharging mode. Lung-Sheng Yang was born in Taiwan, ROC, in 1967. He received his B.S. degree in Electrical Engineering from the National Taiwan Institute of Technology, Taipei, Taiwan, in 1990; his M.S. degree in Electrical Engineering from National Tsing Hua University, Hsinchu, Taiwan, in 1992; and his Ph.D. degree in Electrical Engineering from National Cheng Kung University, Tainan, Taiwan, in 2007. He is presently working as an Assistant Professor in the Department of Electrical Engineering, Far East University, Tainan, Taiwan. His current research interests include power factor correction, dc-dc converters, renewable energy conversion, and electronic ballasts. Chia-Ching Lin was born in Taiwan, ROC, in 1959. He received his B.S. degree from the Department of Electrical Engineering, Far East University, Tainan, Taiwan, in 1980; and his M.S. degree in Electrical Engineering from National Cheng Kung University, Tainan, Taiwan, in 2006. He is presently working as an Assistant Professor in the Department of Electrical Engineering, Far East University. His current research interests include power factor correction and dc-dc converters.