new DCDC power converter is researched for renewable energy and battery hybrid power supplies systems in this paper. At the charging mode, a renewable energy source provides energy to charge a battery via the proposed converter. The operating principle of the proposed converter is the same as the conventional DCDC buck converter. At the discharging mode, the battery releases its energy to the DC bus via the proposed converter. The proposed converter is a nonisolated high stepup DCDC converter. The coupledinductor technique is used to achieve a high stepup voltage gain by adjusting the turns ratio. Moreover, the leakageinductor energies of the primary and secondary windings can be recycled. Thus, the conversion efficiency can be improved. Therefore, only one power converter is utilized at the charging or discharging modes. Finally, a prototype circuit is implemented to verify the performance of the proposed converter.
I. INTRODUCTION
The use of the fossil fuels causes environmental pollution and ecological problems. In addition, carbondioxide emissions result in global warming. Therefore, the development and application of renewable energy sources has become very important
[1]

[10]
. Renewable energies include wind power, solar power, ocean energy, hydrogen energy, hydro energy and so on.
Fig. 1
(a) shows the conventional renewable energy and battery hybrid power supply system. It can be seen that the battery is charged from a renewable energy source via two converters
[11]

[17]
. Therefore, the conventional hybrid power supply system results in energy waste. For this reason, this paper presents a new DCDC converter for threeport renewable energy and battery hybrid power supplies systems, as shown in
Fig. 1
(b). It can be seen that the battery is only via one converter at the charging or discharging mode.
Fig. 2
shows the circuit configuration of the proposed converter. The pulsewidth modulation (PWM) technique is used for the proposed converter. At the charging mode, the renewable energy source provides its energy to charge the battery via the proposed converter. Here the proposed converter is a conventional DCDC buck converter
[18]
,
[19]
. At the discharging mode, the battery releases its energy to the DC bus via the proposed converter. Meanwhile, this converter is a nonisolated high stepup DCDC converter
[20]
. The coupledinductor technique is used to achieve a high stepup voltage gain by adjusting the turns ratio. Moreover, the leakageinductor energies of the primary and secondary windings can be recycled. Thus, only one power converter is utilized at the charging and discharging modes. The operating principles and steadystate analyses will be described in the following sections for the charging and discharging modes.
Renewable energy and battery hybrid power supplies system. (a) Conventional type, (b) Threeport type.
Circuit configuration of proposed DCDC converter.
In order to analyze the steadystate characteristics of the proposed converter, some conditions are assumed: (1) Because the capacitors
C
_{1}
,
C_{B}
, and
C_{bus}
are sufficiently large, the voltages across these capacitors can be treated as constant during each switching period. (2) The ONstate resistance of the switches, the forward voltage drop of the diodes, and the equivalent series resistances of the coupled inductor and capacitors are ignored.
II. CHARGING MODE
When the proposed converter is operated in the charging mode, the primary winding of the coupled inductor is used for a general inductor. The equivalent circuit of the proposed converter at the charging mode is shown in
Fig. 3
, where
R_{B}
is the equivalent load for the battery. The PWM technique is used to control switch
S
_{1}
. Switch
S
_{2}
is used for the synchronous rectifier.
Fig. 4
shows typical waveforms of the proposed converter with the continuous conduction mode (CCM) operation at the charging mode. The operating principles and steadystate analysis are described as follows:
1) Mode 1:
During this time interval [
t
_{0}
,
t
_{1}
], switch
S
_{1}
is turned on and switch
S
_{2}
is turned off. The current flow path is shown in
Fig. 5
(a). The renewableenergy source
V_{in}
supplies its energy to for the coupled inductor
L_{m}
, capacitor
C_{B}
, and load
R_{B}
. Meantime, the coupled inductor
L_{m}
stores its energy. Thus, the current
i_{Lm}
is increased. The voltage across the coupled inductor
L_{m}
is given by:
2) Mode 2:
During this time interval [t
_{1}
, t
_{2}
], switch S
_{1}
is turned off and switch S
_{2}
is turned on. The current flow path is shown in
Fig. 5
(b). Meanwhile, switch S
_{2}
is utilized for a synchronous rectifier. The energy stored in the coupled inductor L
_{m}
is released to the capacitor C
_{B}
and load R
_{B}
. The voltage across the coupled inductor L
_{m}
is derived as:
By using the voltagesecond balance principle on the coupled inductor
L_{m}
, the following equation is obtained:
Substituting (1) and (2) into (3), the voltage gain in the charging mode is found to be:
When the proposed converter is operated in the boundary conduction mode (BCM), typical waveforms are shown in
Fig. 6
. Thus, the peak value of the coupledinductor current is written as:
By utilizing the amperesecond balance principle on the capacitor
C_{B}
, the following equation can be expressed as:
From (6), the peak value of the coupledinductor current can be rewritten as:
Then, the normalized inductor time constant is defined as:
Substituting (4), (5), and (8) into (7), the boundary normalized inductor time constant is given by:
If
τ
_{Lm1}
is larger than
τ
_{Lm1,B}
, the proposed converter is operated in the CCM at the charging mode.
Equivalent circuit of proposed converter at charging mode.
Some typical waveforms of proposed converter with CCM operation at charging mode.
Current flow path of proposed converter at charging mode. (a) Mode 1. (b) Mode 2.
Some waveforms of proposed converter with BCM operation at charging mode.
III. DISCHARGING MODE
The equivalent circuit of the proposed converter at the discharging mode is shown in
Fig. 7
, where
R_{L}
is the equivalent load for the DC bus. The PWM technique is used to control switch
S
_{2}
. Switch
S
_{1}
is turned off at the discharging mode.
Fig. 8
shows typical waveforms of the proposed converter with the CCM operation at the discharging mode. The operating principles and steadystate analysis are described as follows:
1) Mode 1:
During this time interval [
t
_{0}
,
t
_{1}
], switch
S
_{2}
is turned on. The current flow path is shown in
Fig. 9
(a). The energy of the battery is released to the magnetizing inductor
L_{m}
and primary leakage inductor
L
_{k1}
. Thus, the currents
i_{Lm}
and
i
_{Lk1}
are increased. The secondary leakage inductor
L
_{k2}
, secondary winding
N
_{2}
, capacitor
C
_{1}
, and battery are in series to release their energies for the load
R_{L}
. The energy of the capacitor
C_{bus}
is also provided to the load
R_{L}
. Therefore, the current
i
_{Lk2}
decreases. At
i
_{Lk2}
= 0, the energy stored in the leakage inductor
L
_{k2}
is completely recycled to the load
R_{L}
.
2) Mode 2:
During this time interval [t
_{1}
, t
_{2}
], switch S
_{2}
is still turned on. The current flow path is shown in
Fig. 9
(b). The energy of the battery is still released to the magnetizing inductor L
_{m}
and primary leakage inductor L
_{k1}
. The energy of the capacitor C
_{bus}
is provided to the load R
_{L}
.
3) Mode 3:
During this time interval [t
_{2}
, t
_{3}
], switch S
_{2}
is turned off. The current flow path is shown in
Fig. 9
(c). The energies stored in the magnetizing inductor L
_{m}
and primary leakage inductor L
_{k1}
are released the capacitor C
_{1}
. Thus, the currents i
_{Lm}
and i
_{Lk1}
are decreased. The secondary winding N
_{2}
, capacitor C
_{1}
, and battery are in series to release their energies for the leakage inductor L
_{k2}
, capacitor C
_{bus}
, and load R
_{L}
. Therefore, the current i
_{Lk2}
is increased. At i
_{Lk1}
= 0, the energy stored in the leakage inductor L
_{k1}
is completely recycled to the capacitor C
_{1}
.
4) Mode 4:
During this time interval [t
_{3}
, t
_{4}
], switch S
_{2}
is still turned off. The current flow path is shown in
Fig. 9
(d). The secondary leakage inductor L
_{k2}
, secondary winding N
_{2}
, capacitor C
_{1}
, and battery are in series to release their energies for the capacitor C
_{bus}
and load R
_{L}
. Therefore, the currents i
_{Lm}
and i
_{Lk2}
are decreased.
When switch
S
_{2}
is turned on, the following equation can be represented as:
Thus:
where the turns ratio of the coupled inductor
n
=
N
_{2}
/
N
_{1}
, and the coupled coefficient
k
=
L_{m}
/(
L_{m}
+
L
_{k1}
).
From the operating principle, it is known that the energy stored in the primary leakage inductor
L
_{k1}
is recycled to the capacitor
C
_{1}
. By using the amperesecond balance principle on the capacitor
C
_{1}
, the released time duration of the primary leakageinductor energy can be expressed as
[20]
:
By utilizing the voltagesecond balance principle on the primary leakage inductor
L
_{k1}
and magnetizing inductor
L_{m}
, the following equations can be obtained:
where
v
_{Lk1(tr)}
is the voltage across the primary leakage inductor
L
_{k1}
during the time duration [
t
_{2}
,
t
_{3}
] and the voltage
v
_{N1(OFF)}
across the magnetizing inductor
L_{m}
during the time duration [
t
_{2}
,
t
_{4}
].
Substituting (11) and (14) into (15), yields:
By substituting (12) into (16), it is possible to derive:
From
Fig. 9
(c), the following equation can be obtained:
Substituting (17) and (18) into (19), the voltage across the capacitor
C
_{1}
is written as follows:
During the switch
S
_{2}
OFFperiod, the following voltage equation is found as:
Therefore:
Using the voltagesecond balance principle on the secondary winding of the coupled inductor
N
_{2}
, the equation can be represented as:
Substituting (13), (20), and (22) into (23), the voltage gain can be found as follows:
At
k
= 1, equation (24) is rewritten as:
The voltage gain with parasitic components is analyzed as follows. In order to simplify the analysis, the leakage inductors of the coupled inductor are neglected. The equivalent circuit is shown in
Fig. 10
.
r
_{N1}
and
r
_{N2}
represent the equivalent series resistances (ESR) of the primary and secondary windings of the coupled inductor.
V
_{FD2}
and
r
_{D2}
are the ONstate forward voltage drop and resistance of
D
_{2}
.
r
_{S2}
denotes the ONstate resistance of
S
_{2}
.
Equivalent circuit of proposed converter at discharging mode.
Some typical waveforms of proposed converter with CCM operation at discharging mode.
Current flow path of proposed converter at discharging mode. (a) Mode1. (b) Mode 2. (c) Mode 3. (d) Mode 4.
Equivalent circuit including ESR of coupled inductor, ONstate forward voltage drop and resistance of diodes, and ONstate resistance of switch. (a) S_{2} ON. (b) S_{2} OFF.
When switch
S
_{2}
is turned on, the equivalent circuit is shown in
Fig. 10
(a). The average values of
i_{bus}
and
v
_{N1}
are written as:
When switch
S
_{2}
is turned off, the equivalent circuit is shown in
Fig. 10
(b). The average values of
i_{bus}
and
v
_{N1}
are found as:
Since the leakage inductors of the coupled inductor are neglected, the coupling coefficient
k
is equal to 1. From (20), the voltage
V
_{c1}
can be rewritten as:
Substituting (30) into (29), yields:
By using the amperesecond balance principle on
C_{bus}
, the following equations are obtained as:
Substituting (26) an (28) into (32),
I
_{b(OFF)}
is derived as:
In addition,
I
_{b(ON)}
can be given as:
Using the voltsecond balance principle on
L_{m}
, yields:
Substituting (27), (31), (33), and (34) into (35), the actual voltage gain can be obtained as follows:
The curves of the ideal and actual voltage gain under
n
=3,
r
_{N1}
=
r
_{N2}
=100 mΩ,
r
_{D2}
=50 mΩ,
r
_{S2}
=50 mΩ,
V
_{FD2}
=1.25 V,
V_{bat}
=24 V, and
R_{L}
=200 Ω are plotted in
Fig. 11
. It can be seen that the proposed converter in the discharging mode can achieve a high stepup voltage gain.
Ideal and actual voltage gain in discharging mode.
When the proposed converter is operating in the BCM, typical waveforms are shown in
Fig. 12
. Thus, the peak value of the magnetizinginductor current is given as:
Applying the amperesecond balance principle on the capacitor
C_{bus}
, the following equation is found:
From the above equation, the peak value of the magnetizinginductor current is rewritten as:
Then, the normalized inductor time constant is defined as:
Substituting (24), (37), (40) into (39), the boundary inductor time constant is obtained as follows:
At
k
= 1,
τ
_{Lm2}
can be rewritten as:
If
τ
_{Lm2}
is larger than
τ
_{Lm2,B}
, the proposed converter in the discharging mode is operated in the CCM.
Some waveforms of proposed converter with BCM operation at discharging mode.
IV. EXPERIMENTAL RESULTS
In order to verify the feasibility of the proposed converter, a prototype circuit is built for a fuelcell and battery hybrid power supply system. The electric specifications and circuit components are selected as the input voltage
V_{in}
= 28~36.5 V, output power
P_{o}
= 200~20 W, battery voltage
V_{bat}
= 24 V, DCbus voltage
V_{bus}
= 200 V, switching frequency
f_{s}
= 50 kHz, coupled inductor
L_{m}
= 72
μ
H and
n
= 3, capacitors
C_{B}
=
C
_{1}
=
C_{bus}
= 220
μ
F, switches
S
_{1}
(IXFH80N085) and
S
_{2}
(IXTQ96N20P), and diodes
D
_{1}
(DSSK6002AR) and
D
_{2}
(DSEP3006A).
Fig. 13
shows the experimental results at the charging mode. The electrical specifications are
V_{in}
= 28 V,
V_{bat}
= 24 V, and full load
P_{o}
= 200 W. From
Figs. 13
(a) and
13
(c), it can be seen that the current waveforms,
i
_{S1}
and
i_{Lm}
, are same during the
S
_{1}
ONperiod. As can be seen from
Figs. 13
(b) and
13
(c), the current waveforms,
i
_{S2}
and
i_{Lm}
, are the same during the
S
_{1}
OFFperiod.
Fig. 13
(c) shows that the proposed converter is operated in the CCM. The measured efficiency at the charging mode is shown in
Fig. 15
. The measured efficiency is around 95.2%~97.5% under
V_{in}
= 28~36.5 V and
P_{o}
= 200~20 W.
Fig. 14
shows the experimental results at the discharging mode. The electrical specifications are
V_{bat}
= 24 V,
V_{bus}
= 200 V, and full load
P_{o}
= 200 W. The waveforms,
v
_{gs2}
,
v
_{S2}
, and
i
_{S2}
, are shown in
Fig. 14
(a). It can be seen from the waveform
i
_{S2}
that the proposed converter is operated in the CCM. From the waveform
v
_{S2}
, the voltage across
S
_{2}
is clamped at approximately 70 V during the
S
_{2}
OFFperiod. Therefore, a low rated voltage MOSFET can be adopted to reduce the conduction loss. As shown in
Fig. 14
(b), the voltage across
D
_{1}
is clamped at approximately 70 V during the
S
_{2}
OFFperiod. The waveforms,
i
_{D1}
and
i
_{Lk1}
, are shown in
Figs. 14
(b) and
14
(c). It can be seen that they consist with the operating principle. The measured efficiency at the discharging mode is shown in
Fig. 15
. The measured efficiency is around 93.3%~95.4%.
Experimental waveforms of proposed converter at charging mode. (a) v_{gs1}, v_{S1}, i_{S1}, (b) v_{gs2}, v_{S2}, i_{S2}, (c) V_{bat}, i_{Lm}.
Experimental waveforms of proposed converter at discharging mode. (a) v_{gs2}, v_{S2}, i_{S2}, (b) v_{D1}, i_{D1}, (c) V_{bus}, i_{Lk1}.
Measured efficiency of proposed converter.
V. CONCLUSIONS
In the conventional hybrid power supplied system, two power converters are used for the battery charging or discharging modes. The conventional system results in energy waste. A new DCDC converter for renewable energy and battery hybrid power supplied systems is investigated in this paper. At the charging mode, a renewable energy source can provide its energy to charge the battery via the proposed converter. At the discharging mode, the battery can release its energy to the DC bus via the proposed converter. Thus, only one power converter is utilized at the charging or discharging modes. The proposed converter can increase the conversion efficiency. The measured efficiency is around 95.2%~97.5% at the charging mode and it is around 93.3%~95.4% at the discharging mode.
BIO
LungSheng Yang was born in Taiwan, ROC, in 1967. He received his B.S. degree in Electrical Engineering from the National Taiwan Institute of Technology, Taipei, Taiwan, in 1990; his M.S. degree in Electrical Engineering from National Tsing Hua University, Hsinchu, Taiwan, in 1992; and his Ph.D. degree in Electrical Engineering from National Cheng Kung University, Tainan, Taiwan, in 2007. He is presently working as an Assistant Professor in the Department of Electrical Engineering, Far East University, Tainan, Taiwan. His current research interests include power factor correction, dcdc converters, renewable energy conversion, and electronic ballasts.
ChiaChing Lin was born in Taiwan, ROC, in 1959. He received his B.S. degree from the Department of Electrical Engineering, Far East University, Tainan, Taiwan, in 1980; and his M.S. degree in Electrical Engineering from National Cheng Kung University, Tainan, Taiwan, in 2006. He is presently working as an Assistant Professor in the Department of Electrical Engineering, Far East University. His current research interests include power factor correction and dcdc converters.
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