In a gridconnected photovoltaic (PV) system, the traditional Zsource inverter uses a low frequency transformer to ensure galvanic isolation between the grid and the PV system. In order to combine the advantages of both Zsource inverters and transformerless PV inverters, this paper presents a modified singlephase transformerless Zsource PV gridconnected inverter and a corresponding PWM strategy to eliminate the ground leakage current. By utilizing two reversedbiased diodes, the path for the leakage current is blocked during the shootthrough state. Meanwhile, by turning off an additional switch, the PV array is decoupled from the grid during the freewheeling state. In this paper, the operation principle, PWM strategy and commonmode (CM) characteristic of the modified transformerless Zsource inverter are illustrated. Furthermore, the influence of the junction capacitances of the power switches is analyzed in detail. The total losses of the main electrical components are evaluated and compared. Finally, a theoretical analysis is presented and corroborated by experimental results from a 1kW laboratory prototype.
I. INTRODUCTION
For PV gridconnected systems, two types of inverters are usually used. One is a dc/ac inverter with a linefrequency transformer and the other is a dc/ac inverter with a dc/dc converter. This linefrequency transformer can boost the voltage after the dc/ac inverter and guarantee galvanic isolation between the grid and the PV system
[1]
. However, because of its low frequency (50–60Hz), this transformer is big, heavy and expensive
[2]
. Therefore, a high frequency dcdc converter with a highfrequency transformer is used to boost the voltage to reach a constant value
[3]
,
[4]
. Unfortunately, the highfrequency transformer and switches in the dc/dc converter will cause additional power loss. As a result, the interest in singlestage transformerless conversion topologies has grown.
In singlestage topologies, the Zsource inverter is one of the best choices to realize inversion and boost functions in a single stage
[5]

[7]
. It has some advantages such as a simple structure, high reliability of the inverter to avoid the influence of shootthrough due to EMI, and little output waveform distortion. However, the isolation capability has to be considered carefully because of the removal of the transformer. The traditional Zsource inverter topology with its PWM techniques can generate a highfrequency threelevel CM voltage, whose peak value is equal to the Zsource capacitor voltage stress. Because of the capacitance between the PV panel and the ground, the highfrequency potential difference can cause undesirable leakage currents in transformerless PV systems
[8]
. The additional leakage currents can increase the grid current ripples, system losses, and (conducted and radiated) electromagnetic interferences
[9]

[11]
. Thus, a novel transformerless Zsource inverter topology and a particular PWM strategy should be researched to reduce leakage current in order to meet the strict grid codes established by authorities. For example, a 300mA threshold level is stated in the DIN VDE 0216 standard
[12]
.
Recently many research works have been proposed to eliminate the leakage current to meet this standard. These leakage current reduction techniques can be mainly divided into two groups. One is a group of galvanic isolation techniques, and the other is a group of CM voltage clamping techniques.
The galvanic isolation topologies introduce dcdecoupling and acdecoupling methods to disconnect PV systems and the grid during zero states
[13]
. H5 and H6 belong to the dcdecoupling topology family. The H5 topology, which is used in SMA (SMA Solar Technology AG) commercial converters, adds only one switch when compared to fullbridge (FB) inverters
[14]
. The H6 topology, which is proposed in
[15]
, symmetrically adds two additional switches to the FB inverter. The H6 topology equally distributes the device’s efforts and balances the thermal distribution. In
[16]
, a novel H6 topology is proposed, which constructs a new direct power passing path in a half cycle to reduce the conduction loss. The highly efficient reliable inverter concept (HERIC) topology applies a bidirectional switch to realize the disconnection of the converter and the grid during zerovoltage vectors
[17]
. Due to the reduction of switches, the acdecoupling method can provide lower power losses in the conduction path. Although the topologies mentioned above have a simple circuit structure, the galvanic isolation cannot completely eliminate the leakage current due to the influence of switches’ junction capacitances and the parasitic parameters of the leakage current loop
[15]
.
To completely eliminate the leakage current, the CM voltage should be clamped to half of the input voltage during the zero state, which can keep the CM voltage constant for all of the switching modes. This clamping technique has been used in the oH5
[18]
, modified H6 topologies
[19]
, HBZVR
[20]
and HBZVRD
[21]
. By connecting one pole of a PV cell directly to the neutral line of the grid, the leakage current can be eliminated. In
[22]
, the negative pole of a PV array is directly connected to the neutral line of grid. In
[23]
, the positive terminal of a PV array is connected to the phase output during the positive halfwave and to the neutral terminal during the negative halfwave. In addition, the neutral point clamp (NPC) inverter
[24]
connects the midpoint of a PV array to the neutral of a grid, which achieves three or more output levels. However, the NPC inverter, like the halfbridge inverter, demands a higher input voltage.
In this paper, a modified singlephase transformerless Zsource inverter (ZSITL) with one decoupling switch and two fastrecovery diodes is presented to eliminate the ground leakage current. In addition, a special PWM strategy is proposed to avoid zero states with the two lower switches conduction. Furthermore, it can ensure that each phase leg switches on and off once per switching cycle, and it can make the shootthrough zero states evenly allocated into each phase. Moreover, it is analyzed that the junction capacitances of the switches can influence the CV voltage, and a corresponding paralleled capacitor of the switch is designed. Hence, the path for the leakage current can be blocked by two reversebiased diodes during the shootthrough state, while the CM voltage remains constant during the nonshootthrough state. Therefore, the leakage current is eliminated.
The paper is organized as follows. Section II introduces the ZSITL, its PWM strategy and the corresponding operation mode. Section III illustrates the leakage current reduction principle, while Section IV analyzes the system losses. Section V shows some experimental results, and Section VI draws some conclusions.
II. OPERATION PRINCIPLES OF THE MODIFIED TOPOLOGY
 A. Structure of the Modified ZSITL
Fig. 1
shows the ZSITL by using the dcdecoupling method. When compared with the traditional singlephase Zsource gridconnected inverter, the ZSITL adds one additional switch (
S
_{5}
) and one fastrecovery diode (
D
_{2}
).
S
_{5}
is used to electrically decouple the PV array from the grid during the zero state, while
D
_{1}
and
D
_{2}
are used to block the path of the leakage current during the shootthrough state.
Singlephase transformerless Zsource gridconnected PV inverter.
 B. Modified PWM Strategy
The PWM strategy is one of the key factors that affects the leakage current. To achieve the aim of eliminating the leakage current of the ZSITL, the modulation strategy of the traditional singlephase Zsource inverter needs to be modified. For Zsource inverter modulation, the shootthrough states are added to the null intervals to keep the active interval constant. In order to ensure that only one single device is switched during every state transition, the shootthrough states are added adjacent to the instants of the state transitions of a conventional voltagesource inverter. According to the ZSITL topology structure, the zero states where
S
_{2}
or
S
_{4}
conducts must be forbidden to eliminate the leakage current in the modified PWM strategy. Therefore, to achieve the aim of boosting the dclink voltage, the shootthrough states should be added between the active sates and the zero states where
S
_{1}
or
S
_{3}
conducts.
As an illustration,
Fig. 2
shows the switching patterns of the ZSITL. Here,
S
_{1}
is ON and
S
_{2}
is OFF during the positive half cycle. Sinusoidal reference signals are used to modulate
S
_{3}
.
S
_{4}
and
S
_{5}
, with additional shootthrough time intervals added, commutate complementarily to
S
_{3}
. Similarly, during the negative half cycle,
S
_{3}
is ON and
S
_{4}
is OFF.
S
_{1}
commutates at the switching frequency.
S
_{2}
and
S
_{5}
, with additional shootthrough time intervals added, commutate complementarily to
S
_{1}
. It should be noted that each phase leg continues to switch on and off once per switching cycle. Without changing the total zerostate time interval, the shootthrough zero states are evenly allocated into each phase.
Modulation strategy of ZSITL. (a) During the positive half cycle. (b) During the negative half cycle.
Assuming that the inductors
L
_{Z1}
and
L
_{Z2}
have the same inductance (
L
) and that the capacitors
C
_{Z1}
and
C
_{Z2}
have the same capacitance (
C
), the Zsource network becomes symmetrical. As a result, the following formula can be obtained as:
According to the modified PWM strategy, the shootthrough duty ratio
d
_{0}
should be limited to 1
M
.
Therefore, the output peak voltage of the inverter can be expressed as:
where
the peak dclink voltage,
V_{PV}
is the output voltage of the PV panel,
M
is the modulation index, and
T
_{0}
,
T
_{1}
and
T_{S}
are the shootthrough time interval, the nonshootthrough time interval and the switching cycle, respectively. Equation (2) shows that the output voltage can be stepped up or down by choosing an appropriate boost factor,
G
,
If
M
=1
d
_{0}
, the relation between
M
and the maximum value for
G
can be obtained as:
Thus, to boost the output voltage,
M
ranges from 0.5 to 1.
 C. Operation Mode Analysis
Fig. 3
shows the operation modes of the ZVITL.
Six operation modes of ZVITL. (a) Mode 1. (b) Mode 2. (c) Mode 3. (d) Mode 4. (e) Mode 5. (f) Mode 6.

1) Mode 1 is the freewheeling mode during the positive half cycle of the grid voltage.S1andS3are ON whileS2,S4andS5are OFF. The diodesD1andD2are conducting and the Zsource inductor current decreases linearly. The antiparallel diode ofS3conducts. Therefore,uAB=0V and the output current decreases through the switchS1and the antiparallel diode ofS3.

2) Mode 2 is the shootthrough mode during the positive half cycle of the grid voltage.S1,S3,S4andS5are ON whileS2is OFF. The sum of the Zsource capacitors voltage is greater than the dc source voltage (VC1+VC2>VPV), the diodesD1andD2are reversebiased, and the Zsource capacitors charge the Zsource inductors. The Zsource inductor current increases linearly, while the output currentiacdecreases.

3) Mode 3 is the active mode during the positive half cycle of the grid voltage.S1,S4andS5are ON whileS2andS3are OFF. The output current increases through the switchesS5,S1andS4.

4) Mode 4 is the freewheeling mode during the negative half cycle of the grid voltage.S1andS3are ON whileS2,S4andS5are OFF. The diodesD1andD2are conducting. The Zsource inductor current decreases linearly. Therefore,uAB=0V and the output current decreases through the switchS3and the antiparallel diode ofS1.

5) Mode 5 is the shootthrough mode during the negative half cycle of the grid voltage.S1,S2,S3andS5are ON whileS4is OFF. The diodesD1andD2are reversedbiased, and the Zsource capacitors charge the Zsource inductors. The Zsource inductor current increases linearly, while the output current decreases.

6) Mode 6 is the active mode during the negative half cycle of the grid voltage.S2,S3andS5are ON whileS1andS4are OFF. The output current increases through the switchesS5,S3andS2.
From the above analysis, it can be seen that the shootthrough states are evenly inserted between the active states and the traditional zero states. The zero states freewheel only through
S
_{1}
and
S
_{3}
.
III. LEAKAGE CURRENT ANALYSIS OF THE ZSITL
 A. Model of the CM Voltage
Fig. 4
(a) shows the CM model for the ZSITL including the most significant stray elements. The most important stray elements that influence the leakage current dynamics include the stray capacitance between the PV array and the ground
C_{PVg}
and the series impedance between the ground connection points of the inverter and the grid
Z_{pg}
. The leakage current
i
_{cm}
flows through the closedloop path consisting of
C_{PVg}
, the Zsource network, the bridge, the filters (
L
_{ac1}
and
L
_{ac2}
), the utility grid, and
Z_{pg}
. Without
Z_{pg}
considered, the total CM voltage
v_{tcm}
is defined as:
where
v_{AN}
represents the voltage between terminal (A) and terminal (N), and
v_{BN}
represents the voltage between terminal (B) and terminal (N).
Commonmode model for ZSITL. (a) Full model. (b) Simplified model.
From (5), if
L
_{ac1}
=
L
_{ac2}
, the total CM voltage is only relevant to
v_{AN}
and
v_{BN}
. The CM voltage
v_{cm}
can be given by the following:
v_{AN}
and
v_{BN}
are determined by the PWM strategy of the ZSITL. Therefore,
v_{AN}
and
v_{BN}
can be regarded as the controlledvoltage sources connected to the negative terminal of the PV array, namely a squarewave voltage sources with a switching frequency. The simplified circuit of
Fig. 4
(b) is finally obtained.
SW
_{1}
and
SW
_{2}
represent the diodes of the ZSITL.
SW
_{1}
and
SW
_{2}
are ON during the nonshootthrough states, and they are OFF during the shootthrough states.
L
_{12}
is obtained by the following:
In the active modes (Mode 3 and Mode 6), the CM voltage can be expressed as:
where
v_{AN}
=
V_{C}
and
v_{BN}
=
V_{PV}

V_{C}
(Taking Mode 3 as an example.).
In the freewheeling modes (Mode 1 and Mode 4), the CM voltage can be obtained as:
where
In the shootthrough modes (Mode 2 and Mode 5), the sum of the two Zsource capacitors’ voltage is greater than the output voltage of the PV panel. Therefore, the diodes
D
_{1}
and
D
_{2}
are reverse biased.
SW
_{1}
and
SW
_{2}
are OFF so that the path for the leakage current is blocked.
According to the above analysis, because the CM voltage is kept constant during the nonshootthrough states and the discharge path of the CM voltage is blocked during the shootthrough states, the leakage current is avoided.
 B. Influence of the Junction Capacitances
In the active mode, the dc and ac sides of the inverter are directly connected by the filter inductors. The operation states and the commonmode voltage are not affected by the junction capacitance of the switches. In the freewheeling mode, the PV panel is disconnected from the grid by
S
_{5}
, and the CM voltage is affected by the junction capacitances of the switches. In the shootthrough mode, because the path of the leakage current is blocked, the influence of the junction capacitances will not be considered. Therefore, when the inverter commutates from the shootthrough mode to the freewheeling mode, the slope of the voltages
v_{AN}
and
v_{BN}
depends on the junction capacitance of the switches, and the CM voltage
v_{cm}
is accordingly affected.
Taking the commutation from Mode 2 to Mode 1 as an example, there are two stages. The other commutation, from Mode 5 to Mode 4, is similar due to the symmetry of the operation modes.
Stage I:
Fig. 5
shows the transient circuit of the commutation from Mode 2 to Mode 1, where
C
_{D1}
and
C
_{D2}
represent the junction capacitances of the diodes
D
_{1}
and
D
_{2}
,
C
_{1}
~
C
_{5}
represent the junction capacitors of the switches
S
_{1}
~
S
_{5}
and
D
_{3}
represents the antiparallel freewheeling diode of
S
_{3}
. When
S
_{4}
and
S
_{5}
are turned OFF, the two charging or discharging circuits are composed of the junction capacitors
C
_{2}
,
C
_{4}
and
C
_{5}
. According to Kirchhof’s current law, the following current equations can be obtained:
where
i
_{1}
and
i
_{2}
represent the currents of the two charging or discharging circuits;
i
_{D3}
is the current of
D
_{3}
; and
i
_{C2}
,
i
_{C4}
and
i
_{C5}
are the currents of
C
_{2}
,
C
_{4}
and
C
_{5}
, respectively.
Transient circuit of commutation from Mode 2 to Mode 1. (a) Transient circuit. (b) Equivalent circuit from Mode 2 to Mode 1. (c) Equivalent circuit at the end of Stage 1.
From (10) to (12), the formula can be derived as follows:
Assuming that the acquired charge of
C
_{D2}
is equal to the discharged charge of
C
_{D1}
, an equivalent circuit model for the transient state can be obtained in
Fig. 5
(b), where the initial potentials in stage 1 are indicated in the brackets (the node N’ is used as a reference potential). It is obvious that the junction capacitors
C
_{2}
and
C
_{4}
are charged by
C
_{5}
in parallel through the filter inductors
L
_{ac1}
and
L
_{ac2}
. Thus, the voltages
v_{AN}
and
v_{BN}
rises until their values are equal to
V_{PV}
/2, and the transient process on Stage 1 ends.
Fig. 5
(c) shows an equivalent circuit model at the end of Stage 1. Based on the charge conservation, it can be found that:
As a result, the relation of the junction capacitors can be obtained as:
Stage II:
Fig. 6
shows a potential resonant circuit in Mode 1 according to (15). The voltages
v_{AN}
and
v_{BN}
become
V_{PV}
/2 synchronously only if
C
_{5}
=
C
_{2}
+
C
_{4}
at the end of transient state I. Therefore, the CM voltage can still remain
V_{PV}
/2. The modified inverter can operate normally in Mode 1, and the condition for eliminating the leakage current is met as analyzed earlier.
Potential resonant circuit in Mode 1.
IV. POWER LOSS CALCULATION AND ANALYSIS
 A. Conduction Losses Analysis
Assuming that the output current is sinusoidal:
where
i
_{ac}
is the output current, and
I_{CM}
is the peak value of the output current.
The conduction losses of
S
_{1}
can be obtained in the positive half cycle as:
where
V_{CEO}
is the saturation voltage drop,
I_{CN}
is the rated current, and
V_{CEN}
is the collectortoemitter voltage at the rated current. This implies a threshold voltage plus a resistance drop.
In the negative half cycle, the conduction losses of
S
_{1}
during the shootthrough time interval are:
where
I_{L}
is the Zsource inductor current.
Another part of the conduction losses of
S
_{1}
is induced by the conduction of the body diode in the negative half cycle.
where
V_{FO}
is the saturation voltage drop, and
V_{FN}
is the diode voltage drop at the rated current.
Thus, the average conduction losses of
S
_{1}
are:
In a similar way, the average conduction losses of
S
_{2}
can be given as:
where:
According to
Fig. 2
, the conduction losses of
S
_{3}
are equal to those of
S
_{1}
, and the conduction losses of
S
_{4}
are equal to those of
S
_{2}
. In addition, the conduction losses of
S
_{5}
are the sum of
S
_{2}
and
S
_{4}
. Therefore, the conduction losses of the switches are:
 B. Switching Losses Analysis
The turnon losses, turnoff losses and recovery power losses of
S
_{1}
can be calculated from (23) to (25), respectively.
where
f_{S}
is the switching frequency, and
t_{rN}
and
t_{fN}
are the rise and fall times of the switch at a rated current, respectively.
t_{rrN}
is the diodes reverse recovery time, and
I_{rrN}
is the peak reverse recovery current.
Thus, the switching losses of
S
_{1}
can be expressed as follows:
In a similar way, the average switching losses of
S
_{2}
can be given as:
The switching losses of
S
_{3}
are equal to those of
S
_{1}
, and the switching losses of
S
_{4}
are equal to those of
S
_{2}
. In addition, the switch state of
S
_{5}
is the same as that of
S
_{4}
in the positive cycle and it is also the same as that of
S
_{2}
in the negative cycle. However, the voltage across
S
_{5}
is doubled when compared with
S
_{2}
or
S
_{4}
. As a result, the switching losses of
S
_{5}
are quadrupled when compared with
S
_{2}
. The total switching losses of the switches are:
By substituting the parameters from the datasheets of an IRGP4062DPbF
[25]
, the total losses with the change of the switching frequency are calculated. The efficiency evaluation of the ZVITL is shown in
Fig. 7
. The output power of the PV inverter is 1kW.
V_{PV}
=320V,
V_{g}
=220V,
M
=0.775, and
d
_{0}
=0.1. When the switching frequency is low, the switching losses are not the main source of the power losses. However, as the switching frequency increases, the distribution of the switching losses increases gradually and becomes the main source of power losses. A lower switching frequency leads to a high total harmonic distortion (THD) of the output current. Therefore, in view of the quality of the output current and the switching losses, a compromised switching frequency of 10kHz is selected in this paper.
Power switches loss distribution for ZSITL.
V. EXPERIMENTAL RESULTS
A 1kW prototype circuit has been designed and tested to verify the performance of the proposed ZVSTL topology.
The detailed components and parameters are as follows: input voltage,
V_{PV}
=320V; Zsource capacitor voltage,
V_{C}
=360V; Zsource capacitor,
C
_{Z1}
=
C
_{Z2}
=940μF; Zsource inductor,
L
_{Z1}
=
L
_{Z2}
=4mH; filter inductor,
L
_{ac1}
=
L
_{ac2}
=4.5mH; grid voltage,
V_{g}
=220Vac; grid frequency,
f_{g}
=50Hz; switching frequency,
f_{S}
=10kHz; parasitic capacitor,
C_{PVg}
=0.1μF; power switches,
S
_{1}
–
S
_{5}
= IRGP4062DPbF; and junction capacitors of the switches,
C
_{1}
–
C
_{5}
=84pF.
The experimental gating signals in a grid cycle are shown in
Fig. 8
. It can be seen that the experimental gating signals
u
_{s1}
,
u
_{s2}
and
u
_{s5}
agree with the analysis results of the PWM scheme, and that the gating signals of
u
_{s2}
and
u
_{s5}
are synchronized well in the negative half cycle.
Gating signals of switches.
According to the principle of the junction capacitors, one additional capacitor with a value of 84pF should be paralleled to
S
_{5}
. In addition, a capacitor with a values of 82pF is applied in this prototype circuit. Because
Z_{pg}
is very small, it is not considered. The CM voltage and the leakage current waveforms of the ZVITL with a paralleled capacitor in the gridcycle and in the PWM cycle are shown in
Fig. 9
(a) and
Fig. 9
(b). The yellow highlighted sections represent the shootthrough states. From (8) and (9),
v_{AN}
=360V,
v_{BN}
= 40V in Mode 3, and
v_{AN}
=
v_{BN}
=160V in mode 1. By choosing a reasonable value for the paralleled capacitor, 82pF,
v_{AN}
=
v_{BN}
=0.5
V_{PV}
is obtained at the ending point of the transient process from the shootthrough mode to the freewheeling mode. Thus,
v_{cm}
is maintained at approximately
V_{PV}
/2. The leakage current
i
_{cm}
is successfully limited to a very small value that is less than 70mA for the peak value and less than 50mA for the rms value. This complies with the DIN V VDEV 012611 standard.
Experimental waveforms of CM voltage and leakage current with paralleled capacitor. (a) Experimental waveforms in the grid cycle. (b) Experimental waveforms in the PWM cycle.
Fig.10
shows the CM voltage and the leakage current waveforms of the ZVITL without a paralleled capacitor. It can be seen that the leakage current with a paralleled capacitor is less than that without a paralleled capacitor.
Experimental waveforms of CM voltage and leakage current without paralleled capacitor. (a) Experimental waveforms in the grid cycle. (b) Experimental waveforms in the PWM cycle.
Fig. 11
shows the dclink voltage
v_{FF’}
, the collectoremitter voltage
v
_{S5}
of
S
_{5}
, and the Z–source inductor current
i_{L}
waveforms. It is clear that
i_{L}
increases in the shootthrough mode and decreases in the nonshootthrough mode. In the freewheeling modes,
S
_{5}
is OFF and
v
_{S5}
is equal to 200V.
Experimental waveforms of v_{FF’}, v_{S5} and i_{L}.
Fig. 12
shows the grid current and voltage waveforms. The gridconnected current is highly sinusoidal and synchronized with the grid voltage. The experimental efficiency of the Zsource PV gridconnected inverter is shown in
Fig. 13
. The maximum efficiency is 95.15%, including the main circuit, control board, and auxiliary power.
Fig. 14
shows a photograph of the proposed inverter.
Experimental waveforms of v_{g} and i_{ac}.
Measured efficiency of ZVITL.
Photograph of the proposed inverter.
VI. CONCLUSION
A modified transformerless Zsource PV gridconnected inverter has been proposed in this paper. The proposed inverter has the following characteristics: (i) a decoupling switch and two reversedbiased diodes are used to eliminate the leakage current, (ii) a modified PWM strategy is implemented, which ensures a single power device switching per state transition, and retain all of the harmonic benefits of conventional modulation strategies, (iii) no shootthrough issue leads to a greatly enhanced reliability, and (iv) a low ac output current distortion can be achieved because there is no dead time. These factors make the modified Zsource inverter suitable for high efficiency and low leakage current transformerless PV gridconnected applications. Finally, experimental results obtained with a 1 kW hardware prototype verify the effectiveness of the proposed inverter.
Acknowledgements
This research work was supported by National Natural Science Foundation of China (51207032), Power Electronics Science and Education Development Program of Delta Environmental & Educational Foundation (DREK2013003), and Fundamental Research Funds for the Central Universities (Grant No. HIT. NSRIF. 2013019).
BIO
Hongpeng Liu was born in Inner Mongolia, China. He received his B.S. degree in Electrical Engineering from the Harbin University of Science and Technology, Harbin, China, in 2000, and his M.S. and Ph.D. degrees in Electrical Engineering from the Harbin Institute of Technology, Harbin, China, in 2006 and 2011, respectively. In 2011, he joined the Harbin Institute of Technology as an Assistant Professor in the Department of Electrical Engineering. His current research interests include photovoltaic generation, Microgrids, and PWM converter/inverter systems.
Guihua Liu was born in Shandong Province, China. She received her B.S., M.S. and Ph.D. degrees in Electrical Engineering from the Harbin Institute of Technology, Harbin, China, in 2000, 2005 and 2009, respectively. In 2006, she joined the Harbin Institute of Technology as an Assistant Professor in the Department of Electrical Engineering, where she has been an Associate Professor since 2014. Her current research interests include photovoltaic generation, airconditioning control technology and switching power supplies.
Yan Ran was born in Shandong Province, China. He received his B.S. and M.S. degrees in Electrical Engineering from the Harbin Institute of Technology, Weihai, China, in 2011 and 2013, respectively. He is presently working toward his Ph.D. degree in Electrical Engineering at the Harbin Institute of Technology, Harbin, China. His current research interests include photovoltaic generation, Zsource inverters and energy storage.
Gaolin Wang was born in Fujian Province, China. He received his B.S., M.S. and Ph.D. degrees in Electrical Engineering from the Harbin Institute of Technology, Harbin, China, in 2002, 2004 and 2008, respectively. In 2009, he joined the Harbin Institute of Technology, as an Assistant Professor in the Department of Electrical Engineering, where he was an Associate Professor from December 2012 to December 2014, and where he has been a Professor since 2015. His current research interests include permanent magnet synchronous motor drives, position sensorless control of AC motors, and digital control of power converters.
Wei Wang was born in Heilongjiang Province, China. She received her B.S. degree in Automatic Test and Control, her M.S. degree in Electrical Engineering, and her Ph.D. degree in Mechanical Electronic Engineering from the Harbin Institute of Technology, Harbin, China, in 1984, 1990 and 2002, respectively. In 1984, she joined the Harbin Institute of Technology as an Assistant Professor in the Department of Electrical Engineering, where she was an Associate Professor from 1995 to 2003, and where she has been a Professor since 2003. Her current research interests include regenerative energy converter techniques, microgrids, softswitching converters, and lighting electronic technology.
Dianguo Xu was born in Heilongjiang Province, China. He received his B.S. degree in Control Engineering from the Harbin Shipbuilding Engineering Institute, Harbin, China, in 1981, and his M.S. and Ph.D. degrees in Electrical Engineering from the Harbin Institute of Technology (HIT), Harbin, China, in 1984 and 1990, respectively. In 1984, he joined the Harbin Institute of Technology as an Assistant Professor in the Department of Electrical Engineering, where he was an Associate Professor from July 1991 to June 1994, and where he has been a Professor since July 1994. He was the Dean of the School of Electrical Engineering and Automation, HIT, from 2000 to 2010, and was the Assistant President of HIT, from 2010 to 2014. He is now the Vice President of HIT. His current research interests include renewable energy generation technology, power quality mitigation, sensorless vector controlled motor drives, and high performance PMSM servo systems.
Teke A.
,
Latran M. B.
2014
“Review of multifunctional inverter topologies and control schemes used in distributed generation systems,”
Journal of Power Electronics
14
(2)
324 
340
DOI : 10.6113/JPE.2014.14.2.324
Islam M. R.
,
Guo Y. G.
,
Zhu J. G.
2014
“A multilevel mediumvoltage inverter for stepuptransformerless grid connection of photovoltaic power plants,”
IEEE J. Photovoltaics
4
(3)
881 
889
DOI : 10.1109/JPHOTOV.2014.2310295
Amirabadi M.
,
Blakrishnan A.
,
Toliyat H. A.
,
Alexander W. C.
2014
“Highfrequency AClink PV inverter,”
IEEE Trans. Ind Electron.
61
(1)
281 
291
DOI : 10.1109/TIE.2013.2245616
Prasanna P. R.
,
Rathore A. K.
2014
“Analysis, design, and experimental results of a novel softswitching snubberless currentfed halfbridge frontend converterbased PV inverter,”
IEEE Trans. Power Electron.
28
(7)
3219 
3230
DOI : 10.1109/TPEL.2012.2222932
Deng K.
,
Zheng J. Y.
,
Mei J.
2013
“Novel switchedinductor quasiZsource inverter,”
Journal of Power Electronics
14
(1)
11 
21
DOI : 10.6113/JPE.2014.14.1.11
Nguyen M.K
,
Lim Y.C.
,
Chang Y.H.
,
Moon C.J.
2013
“Embedded switchedinductor Zsource inverters,”
Journal of Power Electronics
13
(1)
9 
19
DOI : 10.6113/JPE.2013.13.1.9
Liu Y. S.
,
Ge B. M.
,
AbuRub H.
,
Peng F. Z.
2014
“Overview of space vector modulations for threephase Zsource/quasiZsource inverters,”
IEEE Trans. Power Electron.
29
(4)
2098 
2108
DOI : 10.1109/TPEL.2013.2269539
Bradaschia F.
,
Cavalcanti M. C.
,
Ferraz P. E. P.
,
Neves F. A. S.
,
dos santos E. C.
,
da Silva J. H. G. M.
2010
“Modulation for threephase transformerless Zsource inverter to reduce leakage currents in photovoltaic systems,”
IEEE Trans. Ind Electron.
58
(12)
5385 
5395
DOI : 10.1109/TIE.2011.2116762
Lopez O.
,
Freijedo F. D.
,
Yepes A. G.
,
fernandezComesaa P.
,
Malvar J.
,
Teodorescu R.
,
DovalGansoy J.
2010
“Eliminating ground current in a transformerless photovoltaic application,”
IEEE Trans. Energy Convers.
25
(1)
140 
147
DOI : 10.1109/TEC.2009.2037810
Gu B.
,
Dominic J.
,
Lai J. S
,
Chen C. L
,
Labella T
,
Chen B. F
2013
“High reliability and efficiency singlephase transformerless incverter for gridconnected photovoltaic systems,”
IEEE Trans. Power Electron.
28
(5)
2235 
2245
DOI : 10.1109/TPEL.2012.2214237
Lee S. H.
,
Kim K. T.
,
K J. M.
,
K B. H.
2014
“Singlephase transformerless bidirectional inverter with high efficiency and low leakage current,”
IET Power Electron.
458
7
(2)
DOI : 10.1049/ietpel.2013.0074
Hou C. C.
,
Shih C.C.
,
Cheng P. T.
,
Hava A. M.
2014
“Commonmode voltage reduction pulsewidth modulation techniques for threephase gridconnected converters,”
IEEE Trans. Power Electron.
28
(4)
1971 
1979
DOI : 10.1109/TPEL.2012.2196712
Xiao H. F.
,
Liu X. P.
,
Lan K.
2014
“Zerovoltagetransition fullbridge topologies for transformerless photovoltaic gridconnected inverter,”
IEEE Trans. Ind Electron.
61
(10)
5393 
5401
DOI : 10.1109/TIE.2014.2300044
Saridakis S.
,
Koutroulis E.
,
Blaabjerg F.
2013
“Optimal design of modern transformerless PV inverter topologies,”
IEEE Trans. Energy Convers.
28
(2)
394 
404
DOI : 10.1109/TEC.2013.2252013
Yang B.
,
Li W. H.
,
Gu Y. J
,
Cui W. F.
,
He X. N.
2012
“Improved transformerless inverter with commommode leakage current elimination for a photovoltaic gridconnected power system,”
IEEE Trans. Power Electron.
27
(2)
752 
762
DOI : 10.1109/TPEL.2011.2160359
Zhang L.
,
Su K.
,
Xing Y.
,
Xing M.
2014
H6 transformerless fullbridge PV gridtied inverters,”
IEEE Trans. Power Electron.
29
(3)
1229 
1238
DOI : 10.1109/TPEL.2013.2260178
Xiao H. F.
,
Liu X. P.
,
Lan K.
2014
“Optimised fullbridge transformerless photovoltaic gridconnected inverter with low conduction loss and low leakage current,”
IET Power Electron.
7
(4)
1008 
1015
DOI : 10.1049/ietpel.2013.0404
Xiao H. F.
,
Xie S. J.
,
Chen Y
,
Huang R. H.
2011
“An optimized transformerless photovoltaic gridconnected inverter,”
IEEE Trans. Ind Electron.
58
(5)
1887 
1895
DOI : 10.1109/TIE.2010.2054056
Ji B. J.
,
Wang J. H.
,
Zhao J. F.
2013
“Highefficiency singlephase transformerless PV H6 inverter with hybrid modulation method,”
IEEE Trans. Ind Electron.
60
(5)
2104 
2115
DOI : 10.1109/TIE.2012.2225391
Kerekes T.
,
Teodorescu R.
,
Rodriguez P.
,
Vazquez G.
,
Aldabas E.
2011
“A new highefficiency singlephase transformerless PV inverter topology,”
IEEE Trans. Ind Electron.
58
(1)
184 
191
DOI : 10.1109/TIE.2009.2024092
Freddy T. K. S.
,
Rahim N. A.
,
Hew W. P.
,
Che H. S.
2014
“Comparison and analysis of singlephase transformerless gridconnected PV inverters,”
IEEE Trans. Power Electron.
29
(10)
5358 
5369
DOI : 10.1109/TPEL.2013.2294953
Gu Y. J.
,
Zhao Y.
,
Yang B.
,
Li C. S.
,
He X. N.
2013
“Transformerless inverter with virtual DC bus concept for costeffective gridconnected PV power systems,”
IEEE Trans. Power Electron.
28
(10)
793 
805
DOI : 10.1109/TPEL.2012.2203612
Araujo S. V.
,
Zacharias P.
,
Mallwitz R.
2010
“Highly efficient singlephase transformerless inverters for gridconnected photovoltaic systems,”
IEEE Trans. Ind Electron.
57
(9)
3118 
3128
DOI : 10.1109/TIE.2009.2037654
Wang Y.
,
Li R.
2013
“Novel highefficiency threelevel stackedneutralpointclamped gridtied inverter,”
IEEE Trans. Ind Electron.
60
(9)
3766 
3774
DOI : 10.1109/TIE.2012.2204712
International Rectifier
2013
IRGP4062D Datasheet
http://www.irf.com/productinfo/datasheets/data/irgb4062dpbf.pdf