In this paper, a new hybrid DCDC converter is proposed for electric vehicle 3.3 kW onboard battery charger applications, which can be modulated in a phaseshift manner under a fixed frequency or frequency variation. By integrating a halfbridge (HB) LLC series resonant converter (SRC) into the conventional phaseshift fullbridge (PSFB) converter with a fullbridge rectifier, the proposed converter has many advantages such as a full softswitching range without dutycycle loss, zerocurrentswitching operation of the rectifier diodes, minimized circulating current, reduced filter inductor size, and better utilization of transformers than other hybrid dcdc converters. The feasibility of the proposed converter has been verified by experimental results under an output voltage range of 250420V dc at 3.3 kW.
I. INTRODUCTION
For electric vehicle (EV) applications, an ACDC converter is required for use as a battery charger. In general, the battery charger for EVs consists of a power factor corrector and an isolated DCDC converter that directly recharges the battery in EVs
[1]

[5]
. During the battery recharging process, the output of the DC DC converter varies greatly within a specific voltage range and its operating range becomes very wide. This makes it difficult to optimally design a converter in the sense of power conversion efficiency.
A conventional phaseshift fullbridge (PSFB) converter with a fullbridge rectifier is the preferred DCDC topology for EV onboard battery charger applications, because of its natural zerovoltageswitching (ZVS) operation, low current ripple in the battery charging current, and simple structure and control
[6]

[8]
. However, for wideoutputvoltagerange applications like battery chargers, the conventional PSFB converter cannot obtain an optimal power conversion efficiency due to its unique drawbacks such as a narrow ZVS range, large circulating current, and high voltage stress in the rectifier diodes
[9]

[11]
. In order to improve the performance of the conventional PSFB converter for EV battery charger applications, hybrid dcdc converters with an LLC series resonant converter (SRC) integrated into the PSFB converter have recently been researched
[12]

[14]
. By integrating an LLC SRC into a PSFB converter, the resulting hybrid dcdc converters have many advantages such as a wide ZVS range, reduced size of the filter inductor, and reduced voltage stress. Furthermore, the hybrid converter can work with a reduced operating range during the battery recharging process. Thus, it is possible to obtain a more optimal power conversion efficiency.
This paper proposes a novel hybrid PWMresonant converter with a halfbridge (HB) LLC SRC integrated into a conventional PSFB converter with a fullbridge rectifier (FBR). While the proposed converter has all of the benefit of the hybrid converters in
[12]

[14]
, it has fewer components and better utilization of the transformers when compared with previous hybrid converters. The circuit configuration, operating principle, and analysis of the proposed converter are presented in this paper. Its feasibility is verified with experimental results under an output voltage range of 250420 V dc at 3.3 kW.
II. DESCRIPTION OF THE PROPOSED CONVERTER
A circuit diagram of the proposed converter is shown in
Fig. 1
. As can be seen, it is based on a conventional PSFB converter with a FBR. In addition, a halfbridge (HB) LLC SRC with a centertap rectifier (CTR) is integrated into the conventional converter by sharing the laggingleg switches Q
_{3}
and Q
_{4}
and stacking two rectifiers in series before the output filter stage consisting of
L_{O}
and
C_{O1}
. The laggingleg switches in the conventional PSFB converter fail easily in terms of ZVS operation as the output load decreases
[6]

[8]
. However, the laggingleg switches in the proposed converter can achieve ZVS operation without the effect of the output load condition due to the integrated LLC SRC as documented in
[12]

[14]
. Moreover, the circulating current flowing only in the primary side of the conventional PSFB converter is removed because the primary current is reset to zero level with the voltage ripple of the resonant capacitor
C_{R1}
during the freewheeling phase, as shown in the key operating waveforms of the proposed converter described in
Fig. 2
. In the process, the rectifier diodes
D_{14}
can be turned off with the zerocurrentswitching (ZCS) operation. The other output diodes
D_{a1}
and
D_{a2}
are also turned off with the ZCS due to the resonant operation of the LLC SRC. With these improvements, the proposed converter can achieve a higher power conversion efficiency when compared to the conventional PSFB converter. In addition, it can maintain a high efficiency during the battery recharging process.
Proposed converter for electric vehicle onboard battery charger applications.
Key waveforms of the proposed converter.
 A. Operation Principle
In order to analyze the operation of the proposed converter, several assumptions are made as follows.

1) The switch devices are ideal MOSFETs except for the parasitic capacitors and the internal body diodes;

2) The parasitic capacitors of all the MOSFETs have the same capacitance asCOSS;

3) The rectifier diodes are ideal;

4) The output inductorLOis large enough to be treated as a constant current source during a switching period;

5) The magnetizing inductance ofT1is very large so that the magnetizing current can be ignored;

6) The leakage inductor ofT2is included into the resonan t inductorLR;
The driving signals are the same as those for the conventional PSFB converter and the output is regulated in the phaseshift manner at a fixed frequency, which is also the same as the conventional converter. Referring to
Fig. 2
, there are 14 operating modes in a switching period which can be divided into two half cycles:
t_{0}

t_{7}
(from mode 1 to mode 7) and
t_{7}

t_{14}
(from mode 8 to mode 14). The operation principles of two half cycles are symmetric. Therefore, only the first half cycle is described and operating circuits during the cycle are shown in
Fig. 3
.
Operating circuits of the proposed converter. (a) Mode 1. (b) Mode 2. (c) Mode 3. (d) Mode 4. (e) Mode 5. (f) Mode 6. (g) Mode 7.
Mode 1
[
t_{0}

t_{1}
]: Mode 1 begins when the switches
Q_{1}
and
Q_{3}
are in the ONstate, and it ends when the switch
Q_{1}
is turned off. The output current
I_{Battery}
flows through the diodes
D_{1&3}
and the transformers of
T_{1}
and
T_{2}
. The capacitor
C_{O2}
is charged with the resonant current generated by the resonance between
L_{R}
and
C_{R2}
in the LLC SRC via the diode
D_{a2}
. During this mode, the voltage across the resonant capacitor
C_{R1}
is linearly charged by the load current (or
n_{1}I_{Battery}
) reflected to the primary side and it can be expressed as:
The rectifier output voltage
V_{rec}(t)
is the sum of the voltages of the transformers and it is expressed as:
From eq. (2), it is known that both the PSFB and LLC converters transfer the power required in the output port.
Mode 2
[
t_{1}

t_{2}
]: Mode 2 begins when the switch
Q_{1}
is turned off and it ends when the primary voltage
V_{P1}(t)
reaches the zero level. Since the output current
I_{Battery}
still flows through
D_{1&3}
, the voltages across
Q_{1}
and
Q_{2}
are linearly charged or discharged by the energy stored in the output inductor
L_{O}
as follows:
V_{P1}(t)
decreases linearly to the zero level as
V_{Q2}(t)
and
V_{rec}(t)
also ramps down to the output of the LLC SRC,
V_{O1}
. The equation for
V_{rec}(t)
is the same as Equation (2). The resonant current of the LLC SRC continuously charges
C_{O1}
.
At the end of this mode, the switch
Q_{2}
is turned on with ZVS.
Mode 3
[
t_{2}

t_{3}
]: Mode 3 begins when the switch
Q_{2}
is turned on with ZVS, and it ends when the primary current
i_{P1}(t)
reaches the zero level. During this mode,
V_{P1}(t)
is zero,
i_{P1}(t)
is reset to the zero level with the voltage ripple of
C_{R1}
as in eq. (4), and commutation occurs between the output diodes
D_{1&3}
and
D_{a3}
.
V_{rec}(t)
is the output of the LLC SRC,
V_{O1}
. This means that only the LLC SRC transfers the power required in the output port.
In this mode, the circulating current existing in the conventional PSFB converter is removed, and the output diodes
D_{1&3}
are turned off with ZCS. Consequently, the conduction power loss in the active switches and the switching power loss in the rectifier diodes are reduced during the battery recharging process.
Mode 4
[
t_{3}

t_{4}
]: Mode 4 begins when the commutation between
D_{1&3}
and
D_{a3}
is completed, and it ends when the power transfer by the LLC SRC (from
V_{IN}
to
C_{O1}
) is completed. In this mode, the output power is supplied from the output of the LLC SRC as in Mode 3. At the end of this, the diode
D_{a2}
is turned off with ZCS.
Mode 5
[
t_{4}

t_{5}
]: Mode 5 begins when the power transfer by the LLC SRC is completed, and it ends when the switch
Q_{3}
is turned off. The output power is supplied from the output of the LLC SRC as always. The primary current
i_{P2}(t)
becomes the magnetizing current of the transformer
T_{2}
in this mode.
Mode 6
[
t_{5}

t_{6}
]: Mode 6 begins when
Q_{3}
is turned off, and it ends when
V_{P1}(t)
reaches 
V_{IN}
. During this mode, the voltages across
Q_{3}
and
Q_{4}
are charged or discharged by the energy stored in the magnetizing inductor
L_{m2}
of
T_{2}
as follows:
At the end of this mode, the switch Q4 is turned on with ZVS.
Mode 7
[
t_{6}

t_{7}
]: Mode 7 begins when the switch
Q_{4}
is turned on with ZVS. Since
V_{P1}(t)
is the negative input voltage,
i_{P1}(t)
is decreased as in eq. (6) and the commutation between the output diodes
D_{2&4}
and
D_{a3}
occurs. The resonance between
L_{R}
and
C_{R2}
is progressed in the LLC SRC and the resonant current generated by the resonance starts to flow to
C_{O1}
via
D_{a1}
.
The rectifier output voltage
V_{rec}(t)
is equal to eq. (2), and both the PSFB and LLC converters start to transfer the power required in the output port. This mode ends when the commutation between
D_{2&4}
and
D_{a3}
is completed.
 B. Voltage Conversion Ratio, M
The
V_{rec}(t)
in
Fig. 2
can be simplified as in
Fig. 4
(a) after ignoring the durations of modes 2 and 6 because their durations are very narrow in practice.
Fig. 4
(b) shows
V_{rec}(t)
in the conventional PSFB converter with a FBR. As shown in the figure, the conventional PSFB converter processes only the power transfer within the powering interval defined as the effective dutycycle time,
DT_{S}
. On the other hand, the proposed converter continuously transfers the power required in the output port with the help of the LLC SRC. Due to the difference between these powering concepts, the proposed converter can have a lower turnsratio of
T_{1}
when compared with the conventional PSFB converter as in eq. (7).
Output voltage waveform of rectifier: (a) in the proposed converter, (b) in the conventional PSFB converter, and (c) in the hybrid converters in [12] and [13].
Since the lower turnsratio reduces both the conduction power loss generated in the primary side of the transformers and the voltage stress in the rectifier diodes, it is possible that the proposed converter gets a power conversion efficiency that is better than the conventional PSFB converter for charging the battery.
Since the voltage across
C_{R1}
in the proposed converter is small in practice, consider the
V_{rec}(t)
without the effect of
C_{R1}
in
Fig. 4
(a) for getting the voltage conversion ratio of the proposed converter. The voltage conversion ratio,
M
can be obtained by averaging
V_{rec}(t)
as:
where
D
is the dutycycle that is a function of the phaseshifted time,
T_{Φ}
as
D
=
T_{Φ}
/
T_{S}
.
In eq. (8),
M_{LLC_SRC}
is the gain of the LLC SRC and is defined as:
where:
,
R_{L}
=
V_{O1}
/
I_{battery}
, and
f_{S}
is the switching frequency.
From eq. (8), it is known that the output of the proposed converter can be regulated with three methods: phaseshift modulation at a fixed frequency, frequency modulation at a fixed dutycycle, and hybrid modulation of the dutycycle and frequency. In this paper, the control is focused on only the phaseshift modulation at a fixed frequency, which is the same as the conventional PSFB converter.
Fig. 5
shows the curve of
M_{LLC_SRC}
. If the resonant frequency of the LLC SRC,
f_{R}
is designed with the same magnitude as the switching frequency,
M_{LLC_SRC}
has a gain value of 1.0 as shown in the figure. Then, eq. (8) can be fruther simplified and
M
can be expressed by normalizing the voltage conversion ratio based on the turnsratio
n_{1}
of the PSFB transformer as:
where 0 <
α
(=
n_{2}
/
n_{1}
) ≤ 1.
Gain (M_{LLC_SRC}) curve of LLC SRC [16].
Fig. 6
shows the curve of eq. (10). As can be seen, the gain of the proposed converter is always larger than that of the conventional PSFB converter. From this, it is known once more that the turnsratio of
T_{1}
in the proposed converter can be designed with a smaller value when compared with the conventional converter as in eq. (7).
Voltage conversion ratio of proposed converter when f_{R}=f_{S}.
 C. Transformer Utilization, T.U
Fig. 4
(c) shows the
V_{rec}(t)
of the hybrid converters in
[12]
and
[13]
. As can be seen in the figure, the powering sections of the PSFB converter and the LLC SRC are separated over a switching period. This scheme lowers the utilization of the transformers for the battery charger applications as mentioned in
[15]
. For example, if the converters charge a battery with the CCCV charging strategy, the
T.U
in the converters can be calculated as follows:
On the other hand, the PSFB converter in the proposed converter always receives the help of the LLC SRC in processing the power transfer. In this structure, the
T.U
is calculated as follows:
Consequently, the proposed converter has better utilization of the transformers when compared with the hybrid converters in
[12]
and
[13]
.
 D. Reduction of the Circulating Current
The conventional PSFB converter working in the phaseshift manner has freewheeling phases. During the freewheeling phases, the nonpowering current circulates in the primary side of the transformer. This current is officially called the circulating current, and it is pinpointed as a significant drawback of the conventional PSFB converter operating under wide output voltage range applications like battery chargers. This is because the freewheeling phases are maximized when the battery voltage is at its minimum value, and then the conduction power loss coming from the circulating current is maximized.
On the other hand, the circulating current is minimized in the proposed converter due to the voltage ripple of
C_{R1}
as analyzed in detail in mode 3. Aa a result, the RMS value of the primary current can be reduced and the conduction power loss can be also reduced.
 E. SoftSwitching Characteristics
In the conventional PSFB converter, the ZVS operation of the leadingleg switches is less sensitive to output load conditions due to the large ZVS energy coming from the large output filter inductor. On the other hand, the laggingleg switches fail very easily in terms of the ZVS operation as the output load current decreases. This is because the source of the ZVS energy is the energy stored in the small leakage inductor of
T_{1}
and this energy is very small.
The ZVS operation of the leadingleg switches in the proposed converter is obtained with the same principle as the conventional PSFB converter. Hence, it is less sensitive to the output load condition. Meanwhile, the ZVS of the laggingleg switches (or
Q_{3}
and
Q_{4}
) is achieved with the energy stored in the magnetizing inductor of
T_{2}
as explained in eq. (5). This is the same ZVS principle as LLC SRC. Therefore, the laggingleg switches are always able to operate with ZVS without the effect of the output load.
All of the rectifier diodes in the conventional PSFB converter are under hard switching status. Meanwhile, for the proposed converter, the diodes
D_{1}
and
D_{2}
are turned off with ZCS due to the voltage ripple of
C_{R1}
as explained in mode 3. The diodes
D_{a1}
and
D_{a2}
are also turned off with ZCS due to the resonant process of the LLC SRC.
 F. DutyCycle Loss
For a wide ZVS range, the conventional PSFB converter requires the addition of an external inductor. However, this method greatly increases the commutation time of the primary current and a very large dutycycle loss occurs. To compensate for the increased dutycycle loss, the turnsratio of the transformer (
n
) should be designed with a larger value. As a result, both the current stress in the primary side of the transformer and the voltage stress of the rectifier diodes increase significantly.
Meanwhile, the ZVS range in the proposed converter is extended without adding an external inductor. As a result, the commutation time of the primary current is minimized, and the effective dutycycle
D_{eff}
comes close to the real dutycycle
D
, which is defined as
D
=
T_{Φ}
/
T_{S}
. Consequently, the turnsratio of the PSFB transformer
T_{1}
can be designed to be more beneficial when compared with the conventional PSFB converter.
 G. Filter Requirements
In the proposed converter, the output power is supplied continuously from the input stage over a switching period. Meanwhile, the conventional PSFB converter processes only the required power within the effective dutycycle time due to the freewheeling phases. As a result of this difference, the proposed converter can have a much lower
V_{level}
and a smaller current ripple of the output inductor
ΔI_{Lo}
than the conventional converter, as shown in
Fig. 7
. In other words, the output inductor in the proposed converter can be designed with a much smaller inductance when compared with the conventional converter under the same specifications for the current ripple. Eq. (13) shows how to select a magnetic core for the design of the output inductor. As in the equation, the proposed converter can employ an output inductor with a smaller size due to the smaller inductance.
where
A_{P}
is the windowarea product of the magnetic core,
L
is the required inductance,
I_{Lo}
is the RMS or DC value of the current flowing through the inductor,
ΔB
is the working flux density,
J
is the current density, and
k_{w}
is the winding factor.
Current ripple of output inductor current. (a) Conventional PSFB converter. (b) Proposed converter.
III. EXPERIMENTAL RESULTS
To verify the effectiveness of the proposed converter, a 3.3 kW prototype circuit has been built with the components listed in
Table I
and tested with the battery charging strategy consisting of the constantcurrent and constant voltage modes.
Fig. 8
shows the prototype converter. The prototype circuit has the following specifications:

Input voltage:VIN=385 V

Output voltage:VO=250420 V

Charging curent during CC mode:IBattery=7.85 A

Switching frequency:fS=100 kHz
COMPONENTS LIST
Prototype board for the experiment of the proposed converter.
The resonant tank of LLC SRC was designed with the following procedure. Firstly, in order to guarantee the ZVS operation of the laggingleg switches, the following equation should be satisfied.
Eq. (13) can be represented as follows:
Considering the power handing capacity of the halfbridge LLC SRC in EV battery charger applications, the output voltage of the LLC SRC,
V_{O1}
was set as 200V. In order to minimize the circulating current generated by the LLC SRC, the resonant frequency (
f_{R}
) was designed with the same magnitude as the switching frequency (
f_{S}
). For the LLC SRC, the resonant frequency is the loadindependent frequency to guarantee the ZVS turn on of the laggingleg switches from zero to full load. In addition, the gain of the LLC SRC becomes 1.0 at this frequency. Then, in turn, from eq. (9), it can be noted that
n_{2}
is equal to 1.0. According to equation (14), with the design parameters of
n_{2}
=1.0,
V_{IN}
=385V,
V_{O1}
=200V,
C_{oss}
of IPP60R074C6, and
f_{S}
=100 kHz, the magnetizing inductance of
T_{2}
,
L_{m2}
was designed as 120μH. In designing the LLC SRC,
k
is commonly designed in the range from 4.0 to 6.0 considering the input voltage variation and the conduction power loss. In this paper,
k
was selected as 6.0. Then, the resonant inductance for
L_{R}
is given as 20μH. Finally, the resonant capacitor
C_{R2}
can be designed with the following equation:
For the control of the battery charging based on the CCCV charging strategy, two control loops were used: an inner current control loop and an outer voltage control loop. In addition, two PItype controllers were used to regulate the battery voltage and the charging current. The entire control diagram was digitally implemented using a TMS320F28069.
Fig. 9
shows the key operating waveforms of the proposed converter, while the battery is being charged from 250V to 420V with a constantcurrent of 7.85A. As shown in the figure, all of the measured waveforms follow the theoretical waveforms described in
Fig. 2
well. Moreover, it is confirmed that the proposed converter has reduced the circulating current in the primary side when the primary voltage
V_{P1}(t)
is at the zero level during the battery charging process. It can also be seen that there is a staircase voltage waveform in the output of rectifier,
V_{rec}
in the figure. From this, it is known that the output power is transferred continuously from the input stage to the output stage, and that the size of the output filter inductor can be smaller than that of the conventional PSFB converter.
Operating waveforms (i_{p1}(t), i_{p2}(t), V_{p1}(t) and V_{rec}(t)) of the proposed converter during constantcurrent mode with the charging current of 7.85A: (a) when V_{o}=250V, (b) when V_{o}=300V, (c) when V_{o}=350V, and (d) when V_{o}=420V (100% load condition).
Figs. 10
and
11
show the ZVS waveforms of the laggingleg switches during the battery charging process. As can be seen, the laggingleg switches are successfully turned on with ZVS under the overall load conditions due to the combination with an LLC resonant converter.
ZVS waveforms of laggingleg switches during constantcurrent mode with the charging current of 7.85A: (a) when V_{O}=250V, (b) when V_{O}=300V, (c) when V_{O}=350V, and (d) when V_{O}=420V (100% load condition).
ZVS waveforms of laggingleg switches during constantvoltage mode at 420V: (a) when I_{O}=0.78A(10% load), (b) when I_{O}=1.57A(20% load), and (c) when I_{O}=3.93A(50% load).
Fig. 12
shows the efficiency measured during the battery charging process. As shown in
Fig. 12
, the proposed converter has better performance than the conventional PSFB converter. This is due to the reduced circulating current, wider ZVS range, ZCS operation of all of the rectifier diodes, and lower turnsratio. The proposed converter has a maximum efficiency of 97.74% under the full load condition and it maintains high efficiency during the battery charging process.
Efficiency measured with the battery charging profile with constantcurrent mode and constantvoltage mode: (a) Efficiency during constantcurrent mode with the charging current of 7.85A and (b) efficiency during constantvoltage mode at 420V.
IV. CONCLUSION
In order to improve the performance of the conventional PSFB converter for EV battery charger applications, hybrid type converters have been recently researched. In this paper, a new hybrid DCDC converter with a fullbridge rectifier has been presented. While the proposed converter has all of the benefit mentioned for the hybrid converters in
[12]

[14]
, the proposed converter has fewer components and better utilization of the transformers when compared with the previously presented hybrid converters. A summary of the advantages is given as follows:

1) Full ZVS range of all of the active switches without a dutycycle loss

2) ZCS turnoff operation of the rectifier diodes

3) Minimized circulating current

4) Smaller output filter inductor

5) Better utilization of the transformers for battery charger applications
Experimental results with a 3.3 kW prototype converter show a 97.74% peak efficiency and a high efficiency during the battery charging process. From these results, it can be said that the proposed converter is suitable for EV battery onboard charger applications.
BIO
IlOun Lee received his B.S degree in Electrical and Electronic Engineering from Kyungpook National University, Taegu, Korea, in 2000; his M.S. degree in Electrical Engineering from Seoul National University, Seoul, Korea, in 2002; and his Ph.D. degree from the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea, in 2013. From 2003 to 2008, he was a R&D Engineer in the PDP (Plasma Display Panel) Development Group, Samsung SDI, Korea, where he was involved in circuit and product development. From 2008 to 2013, he was a Senior Engineer in the Power Advanced Development Group, Samsung ElectroMechanics Co. LTD., Korea, where he was involved in the development of EV battery chargers, LED lighting drivers, high efficiency server or network power supplies, and high power density adapters. From 2013 to 2015, he was a Senior Researcher in the Energy Saving Lab., Korea Institute of Energy Research (KIER), Daejeon, Korea, where he carried out research on advanced high reliability microgrids and the development of a simulator system for a 20kW diesel engine generator. In 2015, he joined the School of Electrical and Electronic Engineering, Keimyung University, Daegu, Korea, as an Associate Professor. His current research interests include DCDC converters, power factor correction (PFC) ACDC converters, LED drivers, battery chargers for electric vehicles, digital display power systems, and digital control approaches for DCDC converters.
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