High frequency alternating current (HFAC) has been widely used in a wide range of power distribution systems (PDS) due to its superior performance. A high frequency AC/DC converter plays the role of converting HFAC voltage to DC voltage. In this paper, a new
LCLT
resonant AC/DC converter has been proposed, and an easier control method based on input voltage comparison is presented, without the complicated calculation of the zerocrossing point. Both a low distortion and neartounity power factor can be achieved by the proposed resonant converter and control strategy. The operational principle and steadystate analysis are given for the proposed resonant converter. A simulation model and experimental prototype are implemented with an operation frequency of 25kHz and a rated power of 20W. The simulation and experimental results verify the accuracy of the analysis and the excellent performance of the proposed topology.
I. INTRODUCTION
In consideration of the outstanding performance involving fast dynamic response, high power density and flexible voltage grade, high frequency AC (HFAC) plays an important role in a wide range of power distribution systems (PDS), such as spacecraft applications, telecommunication systems, computer and electronic commercial systems, automotive applications and Microgrids
[1]

[14]
. Traditionally, there are two types of power conversion in HFAC PDS: 1) conversion from DC to HFAC
[8]

[14]
, and 2) conversion from HFAC to DC. When compared with DC/HFAC inverters, the following objectives are significant for HFAC/DC conversion: 1) lighter weight and higher power density; 2) controlled DC output over a wide range; 3) lower electromagnetic interference (EMI); 4) higher reliability; 5) lower distortion, nearsinusoidal input current and closetounity power factor.
C_{o}
nsequently, a number of studies have been conducted from the circuit topology to the control scheme.
It has been shown that using passive components is more effective than the active power factor correction (PFC) techniques for HFAC/DC converter for achieving the objectives of a high input power factor and high efficiency
[15]
. When compared with valley fill and charging pump circuits, the resonant converter is a better PFC choice
[16]
,
[17]
. A number of approaches for the design of resonant AC/DC converters have been proposed in
[18]

[23]
. A resonant rectifier based on thyristor switches has been proposed with an
LCT
resonant network and phase shift modulation (PSM). A closetounity power factor and low input harmonics have been achieved in
[18]
,
[19]
. However，the implementation of PSM is complicated for thyristors. Phase density modulation (PDM) has been proposed for AC/DC converter formed by a resonant network, a bidirectional shunt switch, a transformer synchronous rectifiers and an output filter
[20]
,
[21]
. However, more components and a complicated control limit its application. A resonant topology for an AC/DC converter has been reported in
[24]
,
[25]
, which is constituted by an
LCLT
resonant network and a halfbridge switch. If the
LCLT
resonant tank operates at the resonant frequency, the resonant converter behaves like a current source
[26]

[29]
. Moreover, it is inherently output shortcircuit proof due to the existence of clamped diodes
[25]
. The
LCLT
resonant converter satisfies the need for a constant current characteristic and closetounity power factor, but it is incapable of providing a controllable output. The conventional control of output voltage is to regulate the frontend converter. However, this requires more power components and has a high cost
[27]
. In order to simplify the interleaved circuit, this paper presents a controllable converter topology based on a
LCLT
resonant tank and a bidirectional ac switch. In addition, an easier control scheme is presented for the proposed topology without the usual zerocrossing point detection of resonant current.
A wide scope of output regulation can be achieved by the proposed converter and control scheme with a low distortion input current and a closetounity input power factor. This paper is organized as follows. A detailed description of the proposed
LCLT
resonant converter and its operating principle are given in Section II. A steadystate analysis is performed in Section III with a performance discussion, including the output voltage control, input power factor, and total harmonic distortion (THD) of the input current. Simulation results of the
LCLT
resonant AC/DC converter are presented in Section IV to verify the analysis. Section V provides experimental results and Section VI summarizes the conclusion drawn from the investigation.
II. CIRCUIT DESCRIPTION AND OPERATING PRINCIPLE
The proposed AC/DC converter is shown in
Fig. 1
. It is formed by an
LCLT
resonant network, a bidirectional ac switch and a fullbridge rectifier. The
LCLT
resonant network is comprised by a capacitor C and two inductors
L_{1}
and
L_{2}
, where the inductance of
L_{1}
is close to
L_{2}
. The bidirectional ac switch is comprised by two MOSFET switches
Q_{1}
and
Q_{2}
connected back to back. The backend rectifier is constructed by fullbridge diodes
D_{1}
,
D_{2}
,
D_{3}
, and
D_{4}
and an output filter capacitor
C_{o}
. The input is a HFAC voltage
v_{s}
, and
R_{o}
is the load resistance.
The proposed AC/DC converter.
Traditionally, high frequency sinusoidal voltage
v_{s}
is derived from a high frequency AC bus. An
LCLT
resonant circuit converts the HFAC voltage into a current source when the input frequency is equal to the resonant frequency. The impedance of the
LCLT
resonant tank to the fundamental component is significantly less than the impedance to high order harmonics, which effectively reduces the THD of the input current. Therefore, the harmonics distortion of the input side is negligible under the given load conditions. Meanwhile, the fundamental component of the input current and the input voltage are nearly in the same phase if the inductance of
L_{1}
is close to
L_{2}
[25]
. Therefore, a closetounity power factor can be achieved.
When the resonant components
L_{1}
, C and
L_{2}
are tuned to the input frequency, the resonant tank is viewed as an ideal current source without an internal impedance due to a high impedance to the harmonic components
[25]
,
[26]
. With the predefined resonant components and input voltage
v_{s}
, the resonant current passing through the inductor
L_{2}
has a constant amplitude. The bidirectional ac switch is adopted to control the amount of bidirectional current between the resonant tank and the rectifier. The fullbridge rectifier converts the controllable bidirectional current to a unidirectional current for the load. In the existing control of resonant AC/DC converters, it is necessary to calculate the angle between the input voltage and the current pouring into the rectifier. When compared to the complicated current monitor, an easier control scheme based on an input voltage comparison is presented and analyzed in this paper. By means of regulating the conducting angle of
Q_{1}
and
Q_{2}
, the amount of current feeding the rectifier is adjusted. As a result, the output voltage and output current are controllable. A filter capacitor
C_{o}
is used to filter the output ripples and to provide a constant DC voltage to the load.
The operating waveforms of the proposed AC/DC topology are demonstrated in
Fig. 2
.
i_{L2}
is the current passing through
L_{2}
, and
i_{LL}
is the reference waveform with a phase angle lagging 90
^{o}
behind the input voltage. The phase difference between the resonant current
i_{L2}
and
i_{LL}
is
Φ
.
G_{Q1,Q2}
is the driving signals of the switches
Q_{1}
and
Q_{2}
.
v_{o}
is the voltage across the MOSFET switches.
i_{D1},_{D3}
is the current passing through the diodes
D_{1}
,
D_{3}
; while
i_{D2},_{D4}
is the current passing through the diodes
D_{2}
,
D_{4}
.
β_{1}
is the conducting angle of
D_{1}
,
D_{3}
; and
β_{2}
is the conducting angle of
D_{2}
,
D_{4}
during an half cycle of the input voltage.
α
is the charging angle of the output capacitor
C_{o}
. The driving signals are symmetrical with respect to the zero point of the input voltage. Therefore, it is unnecessary to calculate the angle between the input voltage and the current pouring into the fullbridge rectifier. Each operating cycle contains six intervals. The current path of the different states is shown in
Fig. 3
, and the operational analysis is explained as follows.
The waveforms of the proposed AC/DC conver
Current path in switches Q_{1} , Q_{2} and diodes D_{1}, D_{2}, D_{3}, D_{4} in six intervals in one operating cycle.
Interval I
Interval I begins at
ω_{1}t_{1}=(πα)/2
.
Q_{1}
and
Q_{2}
are turned off. The resonant current
i_{L2}
passes through the diodes
D_{2}
and
D_{3}
to charge the output capacitor
C_{o}
. Since the ripple is filtered by the capacitor
C_{o}
, the output voltage is
U_{o}
with a constant value. The voltage
v_{o}
over the switches
Q_{1}
and
Q_{2}
is equal to –
U_{o}
in this interval.
Interval II
Interval II begins at
ω_{1}t_{2}=π/2Φ
. Since the direction of
i_{L2}
has reversed, the resonant current
i_{L2}
passes through the diodes
D_{1}
and
D_{4}
to charge the capacitor
C_{o}
. The voltage
v_{o}
is equal to
U_{o}
in this interval.
Interval III
Interval III begins at
ω_{1}t_{3}=(π+α)/2
.
Q_{1}
and
Q_{2}
are turned on. The resonant current
i_{L2}
passes through
Q_{1}
and
Q_{2}
. The charging process of the capacitor
C_{o}
is suspended.
C_{o}
feeds the load and the voltage
v_{o}
is equal to zero in this interval.
Interval IV
Interval IV begins at
ω_{1}t_{4}=(3πα)/2
.
Q_{1}
and
Q_{2}
are turned off. The resonant current
i_{L2}
passes through the diodes
D_{1}
and
D_{4}
to charge the output capacitor
C_{o}
. The voltage
v_{o}
is equal to
U_{o}
in this interval.
Interval V
Interval V begins at
ω_{1}t_{5}=3π/2Φ
. Since the direction of
i_{L2}
has reversed, the resonant current
i_{L2}
passes through the diodes
D_{2}
and
D_{3}
to charge the capacitor
C_{o}
. The voltage
v_{o}
is equal to –
U_{o}
in this interval.
Interval VI
Interval VI begins at
ω_{1}t_{6}= (3π+α)/2
.
Q_{1}
and
Q_{2}
are turned on. The resonant current
i_{L2}
passes though
Q_{1}
and
Q_{2}
. The charging process of the capacitor
C_{o}
is suspended.
C_{o}
feeds the load and
v_{o}
is equal to zero in this interval.
After interval VI, a new cycle begins and the same operating principle is performed again. In order to make the output stable, the energy stored in the resonant network is regulated by
Q_{1}
and
Q_{2}
in each operating cycle.
III. STEADYSTATE ANALYSIS
Before the steadystate analysis is conducted, some assumptions are made: 1) all of the components and devices are ideal; 2) the input voltage has a constant amplitude with a fixed frequency; 3) the output DC voltage is ripple free and the output voltage is constant.
The converter can be equivalent to a circuit with two sources as shown in
Fig. 4
(a). One is the input voltage source
v_{s}
, and the other is the equivalent voltage source of the output rectifier
v_{o}
. Based on the superposition theorem, the equivalent circuit shown in
Fig. 4
(a) can be decomposed into two parts. The two parts shown in
Fig. 4
(b) and
Fig. 4
(c) can be calculated and analyzed separately.
Equivalent circuit of the controllable LCLT resonant converter.
Since the input voltage
v_{s}
is ideal and without any harmonic components, the input voltage can be expressed as below.
where,
V_{s}
is the root mean square value (RMS) of the input voltage
v_{s}
.
Due to the ideal input voltage, the following relations of the current can be derived from
Fig. 4
(b).
where,
X_{1}
is the fundamental impedance of the inductor
L_{1}
, and
X_{1}
=
ω_{1}L_{1}
=
ω_{1}L_{2}
=
1/ω_{1}C
.
i’_{L1}
and
i’_{L2}
are also sinusoidal waveforms without any high order harmonics.
The voltage
v_{o}
can be expressed by the Fourier series as below.
where,
U_{o}
is the output voltage, and
A_{n}=cos(nα/2)cos(nΦ)
,
B_{n}=sin(nΦ)
, and
θ_{n}=arctan(B_{n}/A_{n})
. The circuit analysis is performed by a series of harmonics.
The currents of
i"_{L1}
and
i"_{L2}
can be derived from
Fig.4
(c).
 A. Determination of the Phase Difference Angle Φ
Furthermore, the total input current
i_{L1}
can be derived by (2) and (5).
The fundamental component of the input current is:
The angle between the input fundamental current
i_{L1.fundamental}
and the input voltage
v_{s}
is
πθ_{1}
. Furthermore, the input power can be derived as below.
The output power is calculated from the output voltage and the load. The expression of the output power
P_{o}
is given as below.
where,
R_{o}
is the load resistance.
Due to the assumption that all of the components and devices are ideal and without losses, the input power
P_{in}
is equal to the output power
P_{o}
. The following expression can be derived from (9) and (10).
Similarly, the resonant current
i_{L2}
can be derived from (3) and (6).
It can be found from equation (12) that
i_{L2}
can be regulated by the input voltage
v_{s}
and the inductance of
L_{2}
. Since a larger
i_{L2}
leads to more circulating current and reactive power, a suitable selection of
L_{2}
is essential to cut down
i_{L2}
.
As shown in
Fig.2
,
i_{L2}
reaches the zero point at the position
ω_{1}t=(π/2Φ)
. Therefore, the following equation is satisfied.
By putting (11) into (13), the solution of equation (13) can be found. At a given
α
varying from 90
^{o}
to 180
^{o}
, the solution is to find the zero point of a function of
Φ
. As a result:
where,
K=8R_{o}/π^{2}X_{1}
.
The relation curves of
Φ
to the control angle
α
with different value of
K
are shown in
Fig. 5
(a). If the parameter
K
is predefined, the angle
Φ
increases as the control angle
α
increases. The angle
Φ
increases with an increase of
K
. The maximum
Φ
is close to 15
^{o}
when
K
is 1; while the minimum
Φ
is close to 0
^{o}
when
K
is 0.1.
(a) The relation curves of angle Φ to control angle α with different K. (b) The relation curves of voltage U_{o}/B to control angle α at different K. (c) The relation curves of THD to control angle α with different K. (d) The relation curves of THD to Φ with different α. (e) The percentage of third, fifth, seventh harmonic of the input current as a function of control angle α (K=0.1). (f) The relation curves of power factor to control angle α with different K.
 B. Output VoltageUo
The output voltage
U_{o}
can be derived from (11).
where,
α
is the conduction angle of the switches
Q_{1}
and
Q_{2}
. It is adjusted to achieve a desired output.
K
is determined by the resonant components and load condition. The relation curves of the output voltage
U_{o}/B
to control the angle
α
are shown in
Fig. 5
(b). It can be observed that the output voltage Uo increases as
α
grows from 90
^{o}
to 180
^{o}
. The output voltage
U_{o}
can be regulated in a wide range from 29.28% to 99.97% when
K
=0.1. Meanwhile, the range is from 29.16% to 99.22% when
K
=0.5, and it is from 28.78% to 96.98% when
K
=1. The variation scope decreases slowly with the increase of
K
.
 C. THD of the Input Current
The THD of the input current is defined as below.
where,
I_{1}
is the RMS of the fundamental current, and
I_{n}
is the RMS of the n
^{th}
harmonic current. The THD of the input current
i_{L1}
can be derived from (7) and (8) as below.
The relation curves of the THD to control the angle
α
with different values of
K
are shown in
Fig. 5
(c). It can be found that the THD decreases with an increase of the angle
α
. When
K
increases, the THD decreases slightly.
Fig. 5
(d) shows the relation curves of the THD to the angle
Φ
with different values of
α
. When the angle
α
is constant, the denominator of the THD in (17) decreases as the angle
Φ
decreases. This is proportional to the RMS of the fundamental current. Therefore, the THD decreases as the angle
Φ
increases.
Fig. 5
(e) shows the percentage of the third, fifth, seventh and ninth order harmonics of the input current as a function of the control angle
α
at
K=0.1
. It can be found that the third harmonic is the dominant harmonic and that the percentages of the other harmonics are low relatively.
 D. Power Factor
The power factor is defined as below.
where,
θ=πθ_{1}=π arctan(B_{1}/A_{1})
.
The relation curves of the power factor to the control angle
α
with different values of
K
are shown in
Fig. 5
(f). It can be found that as the angle
α
increases from 90
^{o}
to 180
^{o}
, the power factor remains closetounity and increase slightly. When
K
increases, the power factor decreases.
IV. SIMULATION VERIFICATION
A prototype of the proposed
LCLT
resonant converter is simulated with an operation frequency of 25kHz and a rated output power of 20W. The simulation is carried out by PSIM. The input is
v_{s}=48sin(ω_{1}t)V
with an angular frequency
ω_{1}=50000π rad/s
, the MOSFET switches are
IRF530N
with a 100mΩ onstate resistance, the inductance of
L_{1}
and
L_{2}
is 42.5μH, C is a resonant capacitor with 0.94μF and the rated load is
R_{o}
=4.1Ω. The simulation results are shown in
Fig. 6
.
Simulation waveforms of the proposed AC/DC converter, operation frequency is 25 kHz, output voltage is 9V.
The input voltage
v_{s}
and input current
i_{L1}
are demonstrated in
Fig. 6
(a) and
6
(b), respectively. The driving signal of
Q_{1}
and
Q_{2}
is shown in
Fig. 6
(c), the resonant current
i_{L2}
and reference waveforms
i_{LL}
are demonstrated in
Fig. 6
(d), and the output voltage
U_{o}
is shown in
Fig. 6
(e).
It can be found from
Fig. 6
(a) and
6
(b) that both the input voltage
v_{s}
and current
i_{L1}
are sinusoidal and that they have the same phase angle.
Fig. 6
c shows that the driving signal
V_{g}
can be derived by an input voltage comparison. The charging angle of the output capacitor
α
is between 90
^{o}
and 180
^{o}
, which is used to control the output voltage. The inductor current
i_{L2}
is slightly ahead of
i_{LL}
. The calculation results of the PSIM simulation are
PF= 0.996
and
THD=6.2%
. The neartounity power factor and the low THD further verify the accuracy of the analysis and the excellent performance of the proposed resonant HFAC/DC converter.
V. EXPERIMENTAL EVALUATION
A laboratory prototype is designed and tested to validate the characteristics of the proposed HFAC/DC converter. The same parameters are adopted for both the simulation and the experimental evaluations. The voltages are measured by a Probe Master Model 4231, and the current are detected by a Hantek AC/DC Current Clamp CC65.
The operating waveforms are illustrated in
Fig. 7
with a 48sin(ω
_{1}
t)V input voltage and a charging angle of
α
=
120^{o}
. It can be found that the input current
i_{L1}
is sinusoidal and that it has a low harmonics distortion. The input voltage
v_{s}
and
i_{L1}
are almost in the same phase, which ensures a neartounity power factor. The resonant current
i_{L2}
lags behind
v_{s}
by nearly 90
^{o}
, and the angle
Φ
is small. The output voltage
U_{o}
is 9V. The experimental waveforms are in good agreement with the simulation results.
The operating waveforms with 48sin(ω_{1}t)V input voltage and charging angle α=120^{o}. (a) Upper trace: input voltage v_{s}, second trace: input current i_{L1}. (b) Upper trace: input voltage v_{s}, second trace: the driving signal of Q_{1} and Q_{2}. (c) Upper trace: input voltage v_{s}, second trace: the inductor current i_{L2}. (d) Upper trace: input voltage v_{s}, second trace: the output voltage U_{o}.
A varied input can be found in some applications sourced by batteries, fuel cells, etc., and the proposed converter can regulate the output voltage against input variations. When the experimental converter is fed with varied inputs, the typical waveforms are demonstrated in
Fig. 8
and
Fig. 9
with different values of the charging angle
α
.
The operating waveforms with 55sin(ω_{1}t)V input voltage and charging angle α=110^{o}. (a) Upper trace: input voltage v_{s}, second trace: input current i_{L1}. (b) Upper trace: input voltage v_{s}, second trace: the driving signal of Q_{1} and Q_{2}. (c) Upper trace: input voltage v_{s}, second trace: the inductor current i_{L2}. (d) Upper trace: input voltage v_{s}, second trace: the output voltage U_{o}.
The operating waveforms with 40sin(ω_{1}t)V input voltage and charging angle α=130^{o}. (a) Upper trace: input voltage v_{s}, second trace: input current i_{L1}. (b) Upper trace: input voltage v_{s}, second trace: the driving signal of Q_{1} and Q_{2}. (c) Upper trace: input voltage v_{s}, second trace: the inductor current i_{L2}. (d) Upper trace: input voltage v_{s}, second trace: the output voltage U_{o}.
When the input voltage increases to
55sin(ω_{1}t)V
, as shown in
Fig. 8
, the output voltage is kept at 9V with a charging angle of
α
=
110^{o}
. When the input voltage decreases to
40sin(ω_{1}t)V
, as shown in
Fig. 9
, the output voltage is kept at 9V with a charging angle of
α
=
130^{o}
. It can be concluded that the charging angle
α
can be regulated against input variations. In addition, the capability in the presence of input variations is determined by the operational scope of the charging angle
α
and the value of
U_{o}/B
. When the charging angle
α
is operated within the scope of 90
^{o}
to 180
^{o}
,
U_{o}/B
is shown in
Fig. 5
b.
Consequently, the proposed
LCLT
resonant converter and control method are effective for the implementation of a high frequency power supply with precise voltage control. Meanwhile, both a low THD and a neartounity power factor are achieved over the operational scope.
Meanwhile, it can be found from the experimental results that the resonant current
i_{L2}
is about 10A. A larger
i_{L2}
results in design difficulty for the inductors, a larger circulation current and increased power losses. It is significant for parameter design to cut down the resonant current.
The curve of the efficiency to the load and input is illustrated in
Fig. 10
. The power loss mainly comes from the MOSFETs, the resonant components and the rectifier. In view of the large resonant current, the conducting losses of the MOSFETs are high. It can be found in
Fig. 10
that the maximum efficiency is close to 94% at a 2Ω load; while the minimum efficiency is close to 89% at a 6Ω load. As
R_{o}
increases, the control angle
α
decreases to make the output stable. Therefore, the increasing conducting angle of the MOSFETs results in the increasing conducting loss and a lowering of the efficiency. It can also be found that the converter efficiency sourced by
48sin(ω_{1}t)V
is lower than that sourced by
40sin(ω_{1}t)V
. The resonant current increases along with the increasing of
v_{s}
, and the conducting angle of the MOSFETs increases as well. Hence, the conducting loss becomes large and the conversion efficiency become low along with the ascending input voltage.
The curve of efficiency to load and input.
Traditionally, the output voltage of an
LCLT
resonant rectifier is regulated by its frontend converter. The resonant tank is designed for optimal parameters with a maximum efficiency
[30]
. However, the output is uncontrollable for a conventional resonant rectifier since it is incapable of being regulated against input variations. The proposed
LCLT
resonant converter provides an effective output control using bidirectional switches. In addition, a wide scope of output regulation can be accomplished by a simple control strategy. In addition, a low distortion and a closetounity power factor are achieved for input side.
VI. CONCLUSION
In order to improve the load performance of HFAC PDS, a controllable HFAC/DC converter is proposed with the modified
LCLT
resonant tank. Meanwhile, an easier control method is presented for the proposed converter. The output voltage can be effectively regulated by bidirectional switches. The circuit description, operating principles, and steadystate analysis are examined in depth. A low THD and a neartounity power factor are both achieved. A controlled output can be achieved over a wide scope of the charging angle
α
. A simulation schematic and an experimental prototype are implemented with a rated output power of 20W, a rated frequency of 25 kHz, and a rated input peak voltage of 48V. The experimental results are in good agreement with the theoretical analysis and the simulation results. Consequently, the proposed resonant topology and control method are an effective solution for HFAC/DC conversion.
Acknowledgements
The authors gratefully acknowledge the financial support of National Natural Science Foundation of China (No.60904078 and No.51177050), the Fundamental Research Funds for the Central Universities of SCUT (No. D2130830), and Technology R&D Program of Guangzhou (No. 2011J4300024).
BIO
Jun Zeng received her Ph.D. degree in Control Theory and Control Engineering from the South China University of Technology, Guangzhou, China, in 2007. She is an Associate Professor in the School of Electric Power, South China University of Technology. Her current research interests include energy management and intelligence control in distributed generation, and the integration of renewable energy to smart grids.
Xuesheng Li was born in Henan, China, in 1991. He received his B.S. degree in Electrical Engineering and Automation from the Henan Polytechnic University, Jiaozuo, China, in 2013. He is presently working toward his M.S. degree in Power Electronics and Drives at South China University of Technology, Guangzhou, China. His current research interests include resonant converters, high frequency power distribution systems, and renewable energy generation.
Junfeng Liu received his M.S. degree in Control Engineering from the South China University of Technology, Guangzhou, China, in 2005, and his Ph.D. degree from the Hong Kong Polytechnic University, Kowloon, Hong Kong, China, in 2013. From 2005 to 2008, he was a Development Engineer at Guangdong Nortel Network, Guangzhou, China. In 2014, he joined the South China University of Technology, where he is an Associated Professor in the School of Automation Science and Engineering. His current research interests include power electronics applications, nonlinear control, high frequency power distribution systems, and motion control systems.
Strzeleki R.
,
Benysek G.
2008
Power Electronics in Smart Electrical Energy Networks
Springer
175 
201
Jain P.
,
Pinheiro H.
1999
“Hybrid high frequency AC power distribution architecture for telecommunication systems,”
IEEE Transactions on Aerospace and Electronic Systems
35
(1)
138 
147
DOI : 10.1109/7.745687
Liu J.
,
Cheng K. W. E.
2013
“μbased robust controller design of LCLC resonant inverter for highfrequency power distribution system,”
IET Power Electron.
6
(4)
652 
662
DOI : 10.1049/ietpel.2012.0637
Guo W.
,
Jain P. K.
2004
“A Powerfactorcorrected acac inverter topology using a unified controller for highfrequency power distribution architecture,”
IEEE Trans. Ind. Electron.
51
(4)
874 
883
DOI : 10.1109/TIE.2004.831746
Bose B. K.
,
Kin M.H.
,
Kankam M. D.
“High frequency ac vs dc distribution system for next generation hybrid electric vehicle,”
in Proc. IECON
1996
706 
712
Chakraborty S.
,
Simoes M. G.
2009
“Experimental evaluation of active filtering in a singlephase highfrequency ac microgrid,”
IEEE Trans. Energy Convers.
24
(3)
673 
682
DOI : 10.1109/TEC.2009.2015998
Liu J.
,
Cheng K.W.E
,
Ye Y.
2014
“A cascaded multilevel inverter based on switchedcapacitor for highfrequency AC power distribution system,”
IEEE Trans. Power Electron.
29
(8)
4219 
4230
DOI : 10.1109/TPEL.2013.2291514
Luo S.
,
Batarseh I.
2006
“A review of distributed power systems part II: High frequency AC distributed power systems,”
IEEE Aerosp. Electron. Syst. Mag.
21
(6)
5 
14
DOI : 10.1109/MAES.2006.1662037
Jain P.
,
Pahlevaninezhad M.
,
Pan S.
,
Drobnik J.
2014
“A review of high frequency power distribution systems: for space, telecommunication, and computer applications,”
IEEE Trans. Power Electron.
29
(8)
3852 
3863
DOI : 10.1109/TPEL.2013.2291364
Jain P.
,
Tanju M.
1993
“Improved DC/AC interface inverter for highfrequency space applications,”
IEEE Trans. Aerosp. Electron. Syst.
29
(4)
1150 
1163
DOI : 10.1109/7.259518
Liu J.
,
Wai K.
,
Cheng E.
2014
“New power sharing scheme with correlation control for inputparalleloutputseriesbased interleaved resonant inverters,”
IET Power Electron.
7
(5)
1266 
1277
DOI : 10.1049/ietpel.2013.0203
Ye Z.
,
Lam J. C. W.
,
Sen P. C.
2007
“A robust onecycle controlled fullbridge seriesparallel resonant inverter for a highfrequency ac (HFAC) distribution system,”
IEEE Trans. Power Electron.
22
(6)
2331 
2343
DOI : 10.1109/TPEL.2007.909190
Ye Z.
,
Jain P. K.
,
Sen P. C.
2007
“A fullbridge resonant inverter with modified phaseshift modulation for highfrequency ac power distribution systems,”
IEEE Trans. Ind. Electron.
54
(5)
2831 
2845
DOI : 10.1109/TIE.2007.896030
Liu J.
,
Cheng K. W. E.
,
Zeng J.
2014
“A unified phaseshift modulation for optimized synchronization of parallel resonant inverters in high frequency power system,”
IEEE Trans. Ind. Electron.
61
(7)
3232 
3247
DOI : 10.1109/TIE.2013.2279354
Jain P.
,
Tanju M. C.
1997
“A unity power factor resonant AC/DC converter for highfrequency space power distribution system,”
IEEE Trans. Power Electron.
12
(2)
325 
331
DOI : 10.1109/63.558753
Lam J. C. W.
,
Jain P. K.
2008
“A modified valley fill electronic ballast having a current source resonant inverter with improved linecurrent total harmonic distortion (THD), high power factor, and low lamp crest factor,”
IEEE Trans. Ind. Electron.
55
(3)
1147 
1159
DOI : 10.1109/TIE.2007.907668
Qian J.
,
Lee F. C.
,
Yamauchi T.
1999
“An improved charge pump power factor correction electronic ballast,”
IEEE Trans. Power Electron.
14
(6)
1007 
1013
DOI : 10.1109/63.803393
Tanju M. C.
,
Jain P. K.
1994
“Highperformance ac/dc converter for highfrequency power distribution systems: analysis, design considerations, and experimental results,”
IEEE Trans. Power Electron.
9
(3)
275 
280
DOI : 10.1109/63.311260
Jain P.
,
Tanju M. C.
“A unity power factor resonant AC/DC converter for highfrequency space power distribution system,”
in Proc. PESC
1994
3 
9
Musavi F.
,
Jain P. K.
,
Zhang H.
“A resonant AC/DC converter for high frequency power architecture,”
in Proc. INTELEC
2002
497 
503
Lai C.M.
,
Lee R.C.
“A singlestage AC/DC LLC resonant converter,”
in Proc. ICIT
2006
1386 
1390
Musavi F.
,
Master Thesis
2001
“A resonant AC/DC converter for high frequency power distribution systems,”
Concordia University
Canada
Master Thesis
Qiu M.
,
PHD Thesis
2004
“High frequency AC distributed power system for desktop computer applications,”
Concordia University
Canada
PHD Thesis
Luo Q.
,
Zou C.
,
Zhi S.
,
Yan Hua
,
Zhou L.
2013
“A multichannel LED driver based on passive resonant constant current networks,”
inProc. the CSEE
33
(18)
73 
79
Q. Luo
,
S. Zhi
,
Zou C.
,
Lu W.
,
Zhou L.
2014
“An LED driver with dynamic highfrequency sinusoidal bus voltage regulation for multistring applications,”
IEEE Trans. Power Electron.
29
(1)
491 
500
DOI : 10.1109/TPEL.2013.2253335
Pollock H.
1997
“Simple constant frequency constant current loadresonant power supply under variable load conditions,”
Electronics Letters
33
(18)
1505 
1506
DOI : 10.1049/el:19971063
Carbone R.
,
Dommel H. W.
,
Langella R.
,
Testa A.
2002
“Analysis and estimation of truncation errors in modeling complex resonant circuits with the EMTP,”
International Journal of Electrical Power and Energy Systems
24
(4)
295 
304
DOI : 10.1016/S01420615(01)000436
Borage M.
,
Tiwari S.
,
Kotaiah S.
2005
“Analysis and design of an LCLT resonant converter as a constantcurrent power supply,”
IEEE Trans. Ind. Electron.
52
(6)
1547 
1554
DOI : 10.1109/TIE.2005.858729
Borage M.
,
Tiwari S.
,
Kotaiah S.
2007
“LCLT resonant converter with clamp diodes: a novel constantcurrent power supply with inherent constantvoltage limit,”
IEEE Trans. Ind. Electron.
54
(2)
741 
746
DOI : 10.1109/TIE.2007.892254
Frivaldský I. M.
,
Drgoň P.
,
Šáik P.
2013
“Experimental analysis and optimization of key parameters of ZVS mode and its application in the proposed LLC converter designed for distributed power system application,”
International Journal of Electrical Power and Energy Systems
47
448 
456
DOI : 10.1016/j.ijepes.2012.11.016