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Practical Implementation of an Interleaved Boost Converter for Electric Vehicle Applications
Practical Implementation of an Interleaved Boost Converter for Electric Vehicle Applications
Journal of Power Electronics. 2015. Jul, 15(4): 1035-1046
Copyright © 2015, The Korean Institute Of Power Electronics
  • Received : February 10, 2015
  • Accepted : April 01, 2015
  • Published : July 20, 2015
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About the Authors
Huiqing Wen
State Key Laboratory of Electrical Insulation and Power Equipment, Xi’an Jiaotong University, Xi’an, China
Huiqing.Wen@xjtlu.edu.com
Bin Su
Hangzhou Electric Power Bureau, State Grid Corporation of China, Hangzhou, China

Abstract
This study presents a practical implementation of a multi-mode two-phase interleaved boost converter for fuel cell electric vehicle application. The main operating modes, which include two continuous conducting modes and four discontinuous conducting modes, are discussed. The boundaries and transitions among these modes are analyzed with consideration of the inductor parasitic resistance. The safe operational area is analyzed through a comparison of the different operating modes. The output voltage and power characteristics with open-loop or closed-loop operation are also discussed. Key performance parameters, including the DC voltage gain, input ripple current, output ripple voltage, and switch stresses, are presented and supported by simulation and experimental results.
Keywords
I. INTRODUCTION
Fuel cell electric vehicles (FCEVs) have been extensively used nowadays to replace the traditional internal combustion engine-based vehicles because of the advantages of the former in the aspects of high energy density, zero emissions, and high efficiency [1] - [5] . Fuel cell (FC) stacks can provide power for the motors, either stand-alone or combined with renewable energy sources, such as photovoltaic modules or wind turbines [6] . Among various FC technologies available for vehicle application, the proton exchange membrane fuel cell (PEMFC) is regarded as the best candidate because of its high power density and low operating temperatures [7] - [9] . Fig. 1 illustrates a typical volt-ampere characteristic of a 6 kW PEMFC stack, which comprises 65 FCs. The output voltage of the FC stack varies with the stack output current, which is determined with the FCEV output power demand. Fig. 1 shows that the output voltage of the FC stack in the low-power region is higher than that in the high-power region.
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Volt-ampere characteristic of fuel cell stacks based on power demands.
FC stacks exhibit good power capability for steady-state operation; however, their dynamic response is relatively low [10] - [12] . Thus, energy storage systems (ESS), including batteries or supercapacitors, are required to integrate with the FC stacks to provide high transient current and recover energy during the regenerative braking [13] . Fig. 2 shows the FCEV system topology, including the main power source (PEMFC), high power DC-DC converter connecting FC with the high-voltage DC bus, inverter, and motor. An auxiliary power source, including a battery stack or ultracapacitor, is also connected to the high-voltage DC bus via bidirectional DC-DC converter for its rapid dynamic response. The total power of the EV DC bus PDC is provided by two sources, namely, FC stacks PFC and supercapacitors PESD .
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Vehicle system topology of FCEV.
For the FC application, a high-power DC-DC power interface is the key component to achieve excellent power management and full usage of the FC stacks. High voltage ratio is one of the most important requirements for this DC-DC converter. Fig. 1 shows that FC is the main energy source to power the vehicles, and its voltage changes in a wide range. The current ripple and its harmonic content are also determining factors for the FC lifetime [14] , [15] . Volume, cost, weight, manufacturing complexity, and efficiency are also major concerns for the high-power DC-DC converter.
Several topologies have been discussed in prior studies, including conventional boost converter [16] , buck-boost converter [17] , half-bridge [18] or full-bridge converter [19] , and push-pull converter [20] . Considering the increasing demands in power density, non-isolated DC-DC converter is preferred for FCEV application. For the conventional boost converter (CBC), the major concern is the boost inductor because it is the heaviest component in the high-power DC-DC converter. A low inductance is preferred to reduce the inductor size and weight. However, this condition will result in high input current ripple and affect the FC stack lifetime. The CBC efficiency is also lower for a higher duty cycle [21] , [22] . A multi-device structure with interleaved control was proposed in [12] to reduce the input current ripples and the passive component size. However, the component number is doubled, which will lower the system reliability. A high-gain three-level boost converter for FCEV vehicle applications was discussed in [23] . However, the current control for this topology is complicated and the neutral-point voltage balance is still a design challenge. An interleaved reduced-component-count DC-DC converter was discussed in [24] in FCEV with a multi-voltage electric net. However, this system is difficult to use for high power applications.
From the previous analysis, an interleaved boost converter (IBC) can be used to improve the power density and transient response [25] , [26] . For low-power application, the interleaved winding-coupled boost converter is commonly used [27] - [29] . However, for FCEV application, the inductor current reaches 600 A, and the coupled inductor is difficult to manufacture. IBC can be controlled to operate in two modes, namely, continuous conduction mode (CCM) or discontinuous conduction mode (DCM). A total of two CCMs and four DCMs are present for a two-phase IBC. The input current ripple patterns of a two-phase IBC are highly complicated. Thus, discussing the key performance parameters is essential, which include the DC voltage gain, input ripple current, output ripple voltage, and switch stresses that consider the parasitic resistance effects.
In the current study, a practical implementation of a multi-mode two-phase IBC is presented for FCEV applications. Various modes, including two CCMs and four DCMs, are discussed with their equivalent circuits. The effects of the inductor parasitic resistance in the voltage conversion ratio are analyzed. The safe operational area is analyzed through a comparison of the different operation modes. The output voltage and power characteristics with open-loop or closed-loop operation are discussed. Simulation and experimental result are also presented.
II. IBC OPERATING MODES
Fig. 3 depicts the topology of the two-phase IBC. This converter consists of two CBCs interleaved with one IGBT and diodes connected in parallel. The phase-shift interleaved operation will double the inductor current ripple frequency and reduce the size of the inductor. A rapid dynamic response for the converter can also be achieved because of the increased system bandwidth.
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Topology of a two-phase interleaved boost converter.
For the two-phase interleaved IBC, the phase shift between the gate drive signals for Q1 and Q2 is180°. The input current is the sum of the two phase currents. Fig. 1 shows that the operating voltage of the FC stack is changing with different power regions of FCEV. IBC should regulate its duty ratio D to provide a certain voltage to the motor inverter. One constraint is that the input current iin must be designed in CCM because FC is the main power source in the FCEV system. However, each phase current can operate in either CCM or DCM. Thus, a total of two CCMs and four DCMs are present based on the duty ratio range and input current distribution.
Fig. 4 illustrates the typical IBC waveforms for two CCMs. Fig. 4 (a) shows that in the low-power region case, only one inductor current is increasing during 0− δ1 and all switches are off during δ1 π . Thus, during δ1 π , both the inductor and input currents are decreasing. However, Fig. 4 (b) shows that for the high-power region, at least one phase has an increased inductor current for each stage. During 0− δ1 and δ2 π , two inductor currents are increasing for these two stages.
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Typical IBC waveforms with CCM. (a) CCM-I:0<D<0.5; (b)CCM-II:0.5≤D<1.
Fig. 5 illustrates the typical IBC waveforms for DCM. Four operating modes are present based on the duty ratio range and input current distribution. The main differences in these modes are reflected on the boost inductor current waveforms. For DCM, each inductor current has three states, namely, rising, falling, and zero. These three states are symbolized as “R”, “F”, and “Z,” respectively. Thus, considering all the possible combinations of two IBCs, a total of eight operations are possible: “R-R,” “R-F,” “R-Z,” “F-R,” “F-F,” “F-Z,” “Z-R,” and “Z-F.” Fig. 6 illustrates the equivalent circuits. These eight equivalent circuits can explain all the operating modes, whether CCM or DCM. DCM-II is used as an example to analyze the operation briefly.
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Typical IBC waveforms with DCM. (a) DCM-I; (b) DCM-II; (c) DCM-III; and (d) DCM-IV.
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Equivalent circuits of the IBC. (a) “R-R”; (b) “R-F”; (c) “R-Z”; (d) “F-R”; (e) “F-F”; (f) “F-Z”; (g) “Z-R”; (h) “Z-F.”
R-F ” operation during [0− δ1 ]: Q1 is turned on and current iL1 is increasing linearly from zero. Diode D2 is conducting to freewheel the current iL2 . Thus, iL2 is decreasing to zero. Based on the volt-second balance, the duration of the current iL1 decreasing to zero is defined as M and can be expressed as
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where the voltage conversion ratio d = V2 / V1 . The output capacitor current iC during this stage is expressed as
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R-Z ” operation during [ δ1 δ2 ]: Q1 is still on and current iL1 is increasing to the maximum Imax_L1 . Diode D2 is off during this stage. The input current iin is the same as iL1 . The output capacitor current iC is the opposite of the load current io . The current Imax_L1 can be expressed as:
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F-Z ” operation during [ δ2−π ]: Current iL2 remains constantly zero. Diode D1 is conducting to freewheel the current iL1 . Thus, the current iL1 is decreasing linearly. The output capacitor current iC during this stage is expressed as
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A similar operation can be analyzed for the rest stages of “ F-R ,” “ Z-R ,” and “ Z-F .” TABLE I shows that all operating modes can be explained by using the eight possible operations.
OPERATING MODES AND MAIN SUB-CIRCUITS
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OPERATING MODES AND MAIN SUB-CIRCUITS
III. MODES DISTRIBUTION AND BOUNDARIES
- A. DCM and CCM
The input current iin , which is the sum of iL1 and iL2 , can be expressed as
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With the dynamics of the currents iL1 and iL2 shown in Fig. 3 , the analytical expressions of current iin at the switching angles for each operating mode of IBC can be derived. TABLE II shows the expressions of the input current at the switching angles.
EXPRESSIONS OF THE INPUT CURRENT AT THE SWITCHING ANGLES (PU)
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EXPRESSIONS OF THE INPUT CURRENT AT THE SWITCHING ANGLES (PU)
For mode I, the relationship of the voltage conversion ratio d and the duty cycle D can be expressed as
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However, the average current of input current iin for DCM-II can be expressed as
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If the power losses in the components are neglected and the equivalent load resistance is R , then the IBC power transfer characteristic can be expressed as
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Thus,
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Define k = R / Lsfs ; thus, (10) can be simplified as
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As shown in (11), the voltage conversion ratio for DCM-II is affected by the circuit parameters and load conditions. Fig. 7 (a) shows the simulated waveforms of “ iL1 ,” “ iL2 ,” “ iin ,” “ iC ,” and “ vout ” under the condition of “ vin = 100 V, vout = 400 V, Po = 2.91 kW.” IBC operates in CCM-II. However, for the same Po , IBC will operate in DCM-III, as shown in Fig. 7 (b). By using DCM, the inductance will be reduced from 300 μH to 180 μH, and the input current ripple is merely slightly
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Comparison of the simulated waveforms with CCM-II and DCM-III. (a) Simulated inductor current and input current waveforms with CCM-II; (b) Simulated inductor current and input current waveforms with DCM-III; (d) Comparison of the output voltage ripple; (e) Comparison of the output capacitor current.
increased. Fig. 7 (c) shows that the output voltage ripple with DCM-III is lower than that with CCM-II. The reduction of inductance is beneficial for the size reduction and power density improvement. The power devices can also be operated with ZCS; thus, the switching loss of IGBTs and the reverse recovery loss of the diodes can be reduced [30] .
- B. Mode Distribution and Their Boundaries
A total of four criteria determine the mode distribution.
(1) Boundary I : The line of D = 0.5 is the boundary line of CCM-I and CCM-II, DCM-III, and the other three DCMs;
(2) Boundary II : The line D = M will distinguish DCM-I with DCM-II and DCM-IV;
(3) Boundary III : The difference between DCM-II and DCM-IV is the current slope of i L1 and i L2 during [0, δ 1 ]. The boundary line is expressed as
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Combining (1) and (13), the boundary line can be derived
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(4) Boundary IV : For each mode, the total rising and falling duration should be shorter than one switching period. Thus,
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The analytical expressions of the voltage conversion ratio and mode boundaries can be obtained and summarized in TABLE III .
VOLTAGE CONVERSION RATIO,MODE BOUNDARIES, AND INPUT CURRENT RIPPLES
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VOLTAGE CONVERSION RATIO,MODE BOUNDARIES, AND INPUT CURRENT RIPPLES
Based on the boundaries analysis, the modes distributions with their boundaries are shown in Fig. 8 . The boundary line with expression (16) is changing with the voltage conversion ratio d . Expression (1) indicates the relationship between D and M , which is valid for all DCMs. However, the variable pairs ( D , M ) are affected by d , the circuit parameters, and load conditions. Fig. 9 illustrates the mode distribution for various d . Two conclusions can be drawn from Fig. 9 .
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Modes distribution with their boundary lines.
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Modes distribution for various voltage conversion ratios. (a) d=1.2; (b) d=1.5; (c) d=4; (d) d=10.
(1) DCM-I and DCM-II are valid for “ d ≤2.”The DCM-II distribution is wider for a higher d .
(2) DCM-III and DCM-IV are valid for “ d >2.” The DCM-III distribution is wider when the voltage conversion ratio d is larger.
Fig. 10 shows the measured waveforms for the different operating modes. The inductor currents iL1 and iL2 , and input current iin are illustrated and compared. The measured conditions are Vout = 520 V, Vin = 280 V to 320 V, fs = 10 kHz, and Ls = 50 μH. The voltage conversion ratio is less than 2; thus, DCM-I and DCM-II are valid, as shown in Figs. 10 (a) and (b), respectively. Their output power values are 23 kW and 38 kW, respectively. In the test, the duty-cycle ratios are tuned as 38% and 40%. With further increase of the output power, the converter is operating in CCM and the waveforms are shown in Fig. 10 (c). The inductor current is always larger than zero at any instant. The experimental results verified the effectiveness of the previous theoretical analysis.
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Experimental waveforms of inductor current iL1 and iL2 and input current iin with different operating modes: (a) DCM-II with Po = 23 kW; (b) DCM-I with Po = 38 kW; and (c) CCM with Po = 57 kW.
IV. KEY DESIGN ISSUES AND VERIFICATION
- A. Effects of Inductor Parasitic Resistance
Expressions (10) and (11) indicate that the DCM output voltage is determined by both duty-cycle D and circuit parameters k . However, each inductor inevitably has the parasitic resistance ri ( i = 1, 2); thus, the output voltage is also affected. Fig. 6 shows that the state space equations are derived from the equivalent circuits. The IBC steady-state characteristic is expressed as:
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where Dp represents the falling time of the inductor current and is expressed as
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Fig. 11 shows the comparison of the theoretical and experimental measured output voltage ratio d versus the duty-cycle D under the condition of “ k = 40.” The black dotted line represents the ideal voltage conversion ratio. The red dashed line represents the derived d by using expression (17). The experimental results are shown with the blue line with square symbol. Fig. 11 indicates that the experimental measured d is consistent with expression (17). Compared with the ideal result, discrepancy is noted, which is mainly caused by the parasitic resistance of IBC.
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Comparison of the theoretical and experimental results of d versus D.
- B. Output Voltage Ripple
The output voltage ripple is determined by the current flowing through the output capacitor iC , which is the combination of the current iB and the load current io . Fig. 5 illustrates the typical current waveform of iC with DCM-II, which is negative during interval ΔT , and positive during other intervals. Fig. 5 shows that the capacitor current meets the following expression:
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Thus,
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The output voltage ripple is derived as
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For DCM-II, the output voltage ripples can be derived and expressed as
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Similarly, the output voltage ripples with DCM-I can be derived as
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Fig. 12 shows the comparison of calculated input current ripple with simulated and experimental measured results when “ V1 = 320 V, V2 = 520 V, and Co = 600 μF.” Fig. 12 indicates that the expressions derived for the input current ripple are correct regardless of the operating mode. The output power boundary for DCM-I and DCM-II is Po = 55 kW. The inductance in this design is set at 50 μH because the DCM modes are used for the low-power region. The simulation and experimental results indicate that the output voltage ripple ratio is well-regulated within 1%. If only CCMs are used, then the inductance should reach 300 μH. Thus, the hybrid mode shows significant advantages in both keeping low output voltage ripples and improving power density.
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Output voltage ripples versus output power.
- C. Input Current Ripple
Based on the input current expressions presented in TABLE II , the input current ripple with CCM-I is expressed as
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The input current ripple with DCM-I is expressed as
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The average input current with DCM-I is obtained by
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Thus,
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Fig. 13 shows the comparison of the calculated input current ripple with simulated and experimental measured results when “ V1 = 320 V and V2 = 520 V.” Fig. 13 indicates that the expressions derived for the input current ripple are valid for all operating modes. The input ripple ratio for the rated power 150 kW is also well-regulated within 20%.
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Input current ripple. (a) Absolute input current versus output power. (b) Input current ripple versus output power.
- D. Output Power
With DCM, the average output power of IBC can be expressed as
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For the voltage close-loop operation, the voltage conversion ratio d is constant. The load resistance R is decreased with an increasing D to obtain an increased output power. With open-loop operation, for a given V1 and R , Vo and Po will increase with D based on (17) and (30). Fig. 14 shows the relationship of the output power with duty-cycle D for both the open-loop and voltage closed-loop operations. Their base values are the output power values under the condition of “ D = 0.05.” Fig. 14 indicates that the output power will be increased with D for both the open-loop and closed-loop operations. However, with the closed-loop control, the changing speed of the output power is faster than that with the open-loop control.
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Comparison of the output power versus D for the open-loop and closed-loop controls.
- E. SOA Design
Figs. 4 and 5 show the dynamics of the current. Thus, the inductor peak current iLpeak in a switching period can be determined. For CCM, DCM-I, and DCM-III, iLpeak corresponds to iL ( δ1 ); for the other modes, iLpeak corresponds to iL ( δ2 ). The inductor rms current in a switching period Irms can be expressed as:
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Fig. 15 shows the inductor peak current iLpeak and inductor rms current iLrms of IBC as a function of Ls . IGBTs are used as the main power device; thus, the switching frequency is set at 10 kHz. Fig. 15 shows that both iLpeak and iLrms are decreased with the inductance. The inductance determines the output power range. Fig. 15 shows the maximum output power with “ Ls = 100 μH” is 36 kW, while the maximum output power is 68 kW when “ Ls = 50μH.” The comparison results shown in Fig. 7 indicate that the inductance will be significantly reduced by using DCM instead of CCM for the same output power. The reduction of inductance is beneficial for the size reduction and power density improvement. In this design, a hybrid mode is adopted, which considers the entire power range. DCM is used for the low-power region and CCM is adopted for the high-power region. The boundary for DCM and CCM is set at close to “ Po = 70 kW.”
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SOA design of IBC with DCM. (a) Relationship of the inductor peak current iLpeak with Ls; (b) Relationship of the inductor rms current iLrms with Ls.
- F. Efficiency
Fig 16 (a) is the system efficiency curve, which indicates that the entire efficiency is approximately between 95% and 98%; the efficiency is over 97% when the output power is larger than half of the rated power. Fig. 16 (b) shows the estimated power loss distribution of the prototype with Po = 20 kW using the proposed hybrid mode for IBC. DCM is used for the low-power region and CCM is used for the high-power region. The main power losses include the IGBT conduction loss Pcond_IGBT , IGBT switching losses Psw_IGBT , diode conduction loss Pcond_Diode , IGBT switching losses Psw_diode , losses of inductor Pmag_L , and other losses, including the parasitic resistor losses. By using DCM, both the turn-on switching loss of IGBTs and the reverse recovery loss of the diodes can be eliminated. Thus, the efficiency of IBC for the low-power region can be improved from 90% to close to 92%.
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System efficiency curve and power loss breakdown distribution. (a) System efficiency curve; (b) power loss distribution.
V. CONCLUSION
This study presents the operation, design, and performance characteristics of a multi-mode two-phase IBC for FCEV applications. Different modes, including two CCMs and four DCMs, are discussed with their equivalent circuits. Four criteria to determine the mode distribution are also discussed. The mode distributions for various d are illustrated and verified with the experimental results. The effects of the inductor parasitic resistance in the voltage conversion ratio and mode boundaries are analyzed. The safe operational area is analyzed through a comparison of the different operation modes. The output voltage and power characteristics with the open-loop or closed-loop operations are discussed. A hybrid mode is adopted in this design. DCM is used for the low-power region and CCM is adopted for the high-power region. The input inductance is reduced from 300 uH to 50 uH. The input ripple ratio for the rated power 150 kW is well-regulated within 20% and the output voltage ripple ratio is well-regulated within 1%. The system efficiency curve indicates that the entire efficiency is approximately between 95% and 98%; the rated efficiency is over 97%.
Acknowledgements
This study was supported by both the State Key Laboratory of Electrical Insulation and Power Equipment (EIPE15203) and the Jiangsu Province University Natural Science and Research Program (13KJB470013).
BIO
Huiqing Wen obtained a BS and MS in Electrical Engineering from Zhejiang University, Hangzhou, China, in 2002 and 2006, respectively. In 2009, he obtained a PhD in Electrical Engineering from the Chinese Academy of Science, Beijing, China. From 2009 to 2010, he was an electrical engineer working with the GE (China) Research and Development Center Company, Ltd., Shanghai, China. From 2010 to 2011, he was an engineer at the China Coal Research Institute, Beijing, China. From 2011 to 2012, he was a postdoctoral fellow at the Masdar Institute of Science and Technology, Abu Dhabi, United Arab Emirates. He is currently a lecturer at the Xi’an Jiaotong-Liverpool University, Suzhou, China. His research interests include bidirectional DC-DC converter, power electronics in flexible AC transmission (FACTS) applications, electrical vehicles (EVs), and high-power three-level electrical driving system.
Bin Su was born in Wenzhou, China in 1981. He obtained a PhD in Electrical Engineering from Zhejiang University, Hangzhou, China, in 2010. He has authored or co-authored nine published technical papers. His research interests include topologies, modeling, and control in power electronics.
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