An inherent zerovoltage and zerocurrentswitching phaseshifted fullbridge converter with reverseblocking insulatedgate bipolar transistor (IGBT) or nonpunchthrough IGBT is proposed in this paper. This converter not only ensures that the switches in the lagging leg works at zerocurrent switching, but also minimizes circulating conduction loss without any additional auxiliary circuits. A 1.2 kW hardware prototype is designed, fabricated, and tested to verify the proposed topology. The control loop design procedures with smallsignal models are also presented. A simple, lowcost, and robust democratic currentsharing circuit is also introduced and verified in this study. The proposed converter is a suitable alternative for compact, costeffective applications with highvoltage input.
I. INTRODUCTION
Galvanic isolated fullbridge (FB) converter is the standard topology in medium and highpower applications, such as telecom power supplies, Xray generators, electrical vehicles, and traction applications
[1]

[4]
. The main concerns in these fields are reliability, efficiency, power density, cost, and other specific specifications (e.g., wide softswitching range, low topologycomplexity, small circulating current, and minimized duty cycle loss)
[1]

[3]
. Three converter types can fulfill these demands: resonant FB
[3]
, phaseshift pulsewidthmodulated (PS PWM) FB
[1]
, and hybrid resonant and PS PWM FB converters
[5]
,
[6]
.
Resonant circuit topologies, especially variable frequency LLC converters, have become popular in recent years. The major advantages of these topologies are zerovoltage switching (ZVS) or zerovoltage transition and nearly zerocurrent switching (ZCS) for primary switches, ZCS for output diodes, and eliminated output choke. In addition, a wide range of soft switching is achieved even with noload condition. However, the extremely high runaway frequency at noload or shortcircuit condition is a potential threat to system reliability.
Alternatively, additional series inductors are often inserted but can be bulky with more duty cycle loss and circulating current to extend the ZVS range of the classic constantfrequency PS PWM FB converters
[1]

[4]
. Hybrid resonant and PS PWM FB converters significantly lower series inductance with true fullrange soft switching and negligible duty cycle loss features
[5]
,
[6]
. These converters are suitable candidates for electric vehicle chargers at the cost of complicated structures and control strategies.
Highvoltage insulatedgate bipolar transistors (IGBTs) with constant voltage drop are often preferred in threephase 380 Vac/440 Vac or 750 Vdc/1500 Vdc input systems. A series of zerovoltage and zerocurrentswitching (ZVZCS) techniques has been proposed to solve the IGBT tail current issue in the lagging leg (e.g., primary impedance blocking, primary resetting, secondary voltage clamping, and output voltage resetting)
[7]

[12]
. These auxiliary circuits are almost inevitable, and negative effects should also be considered (e.g., steadystate primary current overshoot and additional highvoltage stress of the rectifier during the startup period).
These techniques were recently reviewed and reexamined to achieve the balance between performance and cost with novel commercial SiC and Si devices
[13]
,
[14]
. Complex threelevel FB converters that use lowvoltage MOSFETs are another possible solution
[15]
. New highspeed generations of IGBTs have already been recognized as a costeffective alternative to super junction MOSFETs in zerovoltage transition PS FB highvoltage to lowvoltage DC/DC converters. The capacitive snubber or resonant inductor can be optimized for particular operating points, but not for required wide operating ranges as shown in
[14]
. Measures to improve the efficiency must be carefully selected to avoid conditions wherein a loss mechanism is lowered or partly avoided, whereas others are unintentionally increased, thus canceling the expected benefits. The analysis in
[14]
also shows that the best converter efficiency can be achieved without additional components in the case of IGBT_H3.
Among these nextgeneration IGBTs, ReverseBlocking (RB) IGBTs have been investigated and tested in currentsource inverters, resonant inverters, Ttype neutralpointclamped converters, and matrix AC/AC choppers. RB IGBTs offer more advantages over functionally comparable conventional circuits, such as loss reduction, compact structure, and lower cost
[16]

[22]
.
The present study attempts to determine a low topologycomplexity ZVZCS PS PWM FB converter with few negative effects for highinput voltage application. A novel, inherent ZVZCS PS PWM FB converter without additional auxiliary circuits is also proposed, described, designed, and tested. The proposed converter can achieve ZCS for laggingleg switches without a circulating current with the help of RB IGBT or nonpunchthrough IGBT with RB feature.
II. BASIC OPERATION PRINCIPLE OF THE NOVEL CONVERTER
Fig. 1
illustrates the circuit diagram of the proposed inherent ZVZCS PS PWM FB converter, which consists of the following four parts:

1) the leading leg, including two IGBTsS1andS2with their antiparallel diodesDs1,Ds2;

2) the laggingleg, including two RB IGBTsS3andS4without antiparallel diodes;

3) blocking capacitorCB, main transformerTr, and its leakage inductorLlk;

4) output rectifiersD1,D2, andLCfilterLf,C0.
Circuit diagram of the proposed inherent ZVZCS PS PWM FB converter.
No additional auxiliary ZVZCS circuits are used in the circuit.
The topology operation principles can also be explained by the gate sequences and associated key voltage and current waveforms illustrated in
Figs. 1
and
2
, where
Cs
_{1}
and
Cs
_{2}
are the equivalent capacitance of the IGBTs
S
_{1}
and
S
_{2}
respectively,
v
_{ge_lag}
and
v
_{ge_lead}
are the IGBT drive signals respectively,
V
_{in}
is the input voltage,
V
_{0}
is the output voltage,
v
_{AB}
is FB leg middlepoint voltage,
v
_{c}
is the voltage across the blocking capacitor
C
_{B}
,
v
_{rec}
is the rectifying voltage before the output filter,
i
_{P}
is the main transformer,
T
_{r}
is the primary current,
I
_{L}
is the current through the choke
L
_{f}
,
D
_{eff}
is the effective duty ratio, and
T
_{s}
is the switching period.
Key operation waveforms of the proposed converter.
The topology in the half PWM cycle has six distinct operation modes, as shown
Fig. 3
. Similar operation principles in the second half PWM cycle are not provided because of the symmetric circuit structure. The following assumptions are made at this point to simplify the analysis.

1) All power devices and components are ideal.

2) The output choke is sufficiently large to be treated as a constant current source during a switching period.

3)Cs1=Cs2=Cr.
Operation modes of the proposed converter.
Mode [t_{0}–t_{1}] [Fig. 3(a)]
:
S
_{1}
and
S
_{4}
conduct while
S
_{2}
and
S
_{3}
are both deactivated. The input power is delivered from the primary to the output. The primary current
i
_{p}
=
I
_{L}
/
n
charges the blocking capacitor
C
_{B}
at the same time, where
n
is the main transformer primarytosecondary ratio.
Mode [t_{1}–t_{2}] [Fig. 3(b)]
:
S
_{1}
is turned off, whereas
S
_{4}
conducts at
t
_{1}
. The primary
i
_{p}
(i.e., reflected load current to the primary) charges
C
_{s1}
and discharges
C
_{s2}
linearly. The capacitors provide the ZVS condition for
S
_{1}
as follows:
Mode [t_{2}–t_{3}] [Fig. 3(c)]
: The primary ip fully discharges
C
_{s2}
at
t
_{2}
, and the body or external diode
D
_{S2}
of
S
_{2}
is naturally turned on. Thus,
S
_{2}
can turn on at the zerovoltage condition during this interval.
v
_{AB}
is clamped to zero because of the simultaneous conducting of
D
_{S2}
and
S
_{4}
. Therefore, the blocking capacitor voltage
v
_{c}
decreases the primary current
i
_{p}
.
Given that the reflected secondary
i
_{p}
cannot supply the constant inductor current
i
_{L}
, the secondary rectifier diodes
D
_{1}
and
D
_{2}
both conduct for the freewheeling
i
_{L}
.
Mode [t_{3}–t_{4}] [Fig. 3(d)]
: The primary current reaches zero at
t
_{3}
. Given the RB IGBT,
S
_{4}
does not provide
i
_{p}
the reverse current path,
i
_{p}
is maintained at zero state during this interval, and a circulating current state occurs for the conventional ZVS PS FB. The zero state also provides the ZCS condition for
S
_{4}
to be turned off.
The rectifier diodes
D
_{1}
and
D
_{2}
still conduct and share the load current in the secondary circuit.
Mode [t_{4}–t_{5}] [Fig. 3(e)]
:
S
_{4}
is turned off at the zerocurrent condition at
t
_{4}
. After a short delay of dead time,
S
_{3}
can turn on at
t
_{5}
.
Mode [t_{5}–t_{6}] [Fig. 3(f)]
:
S
_{3}
is turned on by the PWM command at
t
_{5}
.
S
_{3}
is turned on at the zerocurrent condition because of the leakage inductor
L
_{lk}
that limits the increase in primary current
i
_{p}
at the negative direction.
The increasing primary current
i
_{p}
cannot supply the load current during this interval, and both secondary rectifier diodes conduct, which clamps the voltage across the transformer windings at zero.
The primary current
i
_{p}
reaches the reflected load current to the primary at
t
_{6}
, and the input voltage source starts to deliver power from the primary to the output such that the second halfcycle starts at
t
_{0}
.
III. DESIGN CONSIDERATIONS
 A. ZVS Range of the Leading Leg
The ZVS transition of the leading leg is supported by the secondary side filter inductance
L
_{f}
and the transformer leakage inductance
L
_{lk}
. Thus, the ZVS range of the leading leg is relatively wide but only limited at light loads, as illustrated below:
 B. ZCS Range of the Lagging Leg
The ZCS transition of the lagging leg is determined as [
t
_{2}
,
t
_{3}
] and [
t
_{3}
,
t
_{4}
], as shown in
Fig. 2
, where the primary current must decrease to zero before the PWM signal is applied to the IGBT in the lagging leg at
t
_{23}
.
Consider the following:
where
D
_{eff}
is defined as the effective duty ratio shown in
Fig. 2
, and
T
_{s}
is the switching period. Thus, Eq. (3) can be revised as follows:
This condition indicates that
t
_{23}
is independent of the load current and is inversely proportional to
D
_{eff}
. With sufficient
D
_{eff}
that to fulfills the output voltage regulation, the ZCS transition of the lagging leg can be achieved in the total line input and output load range, including the no load condition.
 C. Circulating Current Elimination
Mode [
t
_{3}
–
t
_{4}
] and
Fig. 2
show that the primary current reaches zero at
t
_{3}
and remains at zero because of the adopted RB IGBTs in the lagging leg, which do not provide
i
_{p}
to the reverse current path. Therefore, the circulating current does not exit and helps the efficiency improvement.
 D. IGBT Selection in the Lagging Leg
Currently, the primary manufacturers of RB IGBTs are Fuji, IXYS, Mitsubishi, and Infineon. These manufacturers all have their own design, so the RBIGBT architecture depends on the manufacturer. The architecture of an RBIGBT from IXYS is described in
[24]
. This company modified an NPTIGBT by using isolation diffusion and folding up the lower p
^{+}
layer at the chip edge, as shown in
Fig. 4
. Performing the p
^{+}
n
^{˗}
junction that blocks the reverse voltage prevents breakthrough at the chip edge. The p
^{+}
n
^{˗}
junction continues to the isolation layer at the gate connection. These modifications enable the IGBT to block negative collectoremitter voltages as a normal pn diode and still have the operational behavior of a normal NPTIGBT. The maximum RB voltage for this device is 1200 V.
Architecture of an nonpunchthrough(NPT) IGBT (left) and IXYS (right). RBIGBT with an intrinsic diode (right).
Given the limited RBIGBTs provided in the manufacturers’ product category, mass production and cost issues are also concerns for the proposed novel FB converter. The detailed architecture of an IGBT is reexamined at this point. The bodydrift region junction in
Fig. 5
is the junction that blocks the forward voltage when the device is off, and the junction between the p
^{+}
injection and n
^{˗}
layers is the junction that blocks the reverse voltage. Thus, the NPTIGBT can theoretically block a reverse voltage as high as the magnitude of the forward voltage. The NPTIGBT without a body diode is a possible lowcost solution to replace the RBIGBT for mass production. The optimization of switching performance of the RBIGBT is no longer a key issue that makes the RB IGBT still relatively unacceptable in real applications
[23]
. Common IGBT drivers are sufficient, and the prototype only uses a small driver transformer to drive the NPTIGBT.
Detailed architecture of an IGBT.
 E. Current Sharing Strategy with Multiple Modules
The paralleling of standardized converter modules generally offers several advantages, such as redundancy implementation, expandability of output power, and ease of maintenance. When multiconverter modules operate in parallel, the major issue is loadcurrent sharing among the different modules
[25]
. Among the different approaches, the democratic currentsharing method is preferred for its autonomous currentsharing feature. A simple, lowcost, and robust democratic currentsharing circuit is introduced at this point with diodes, as shown in
Fig. 6
. The connecting current bus after the maximum value detection circuit forces the current reference to be the same, which follows the maximum value of the different voltage loop output. The different inner current loop further regulates the module output current independently.
Currentsharing circuit.
IV. CONTROL LOOP DESIGN FOR VOLTAGE AND CURRENT REGULATIONS
The proposed ZVZCS FB converter is used as a downstream main circuit of a marine leadacid battery charger whose frontend converter is a threephase passive rectifier. The constant current (CC) and constant voltage (CV) charge modes are preferred for a leadacid battery
[26]
. Therefore, the control loops for the voltage and current regulations should be carefully designed
[27]
,
[28]
.
The battery model is complicated because of its electrochemical feature under charge/discharge management. One approach is to model the battery as an equivalent resistor in charging mode, while another approach models the battery as a DC source with its series resistor. Major loop design differences between these models occur at the lowfrequency stage. Key factors such as crossover frequency are unaffected
[29]
. Consider that an electronic loadbased battery emulator is used in this study for convenience. Thus, the leadacid battery is modeled as an equivalent resistor.
Finally, the real leadacid battery is further used to reexamine the controller.
Fig. 7
illustrates that several smallsignal transfer functions are defined as follows:
Current loop gain before compensation:
Current loop gain after compensation:
Modulator can be modeled by a constant gain:
where
V
_{pp}
= 2.35 V is the peaktopeak voltage of the triangular carrier signal.
Control loop block.
G
_{vd}
(
s
) is the dutyratiotooutputvoltage transfer function.
G
_{iLd}
(
s
) is the dutyratiotoinductorcurrent transfer function shown in Eq. (12):
G_{i}
(
s
) is the innerloop compensation gain:
G_{v}
(
s
) is the outerloop compensation gain:
K_{v}
(
s
) = 0.1 is the output voltage sense gain,
K_{i}
(
s
) = 0.1 is the current sense gain,
D
is the FB converter duty ratio,
L
_{f}
is the inductance,
C
_{0}
is the output capacitor,
R
is the load resistance,
R
_{C}
is the equivalent series resistance of the output capacitor, and
R
_{L}
is the equivalent series resistance of the inductor in these transfer functions.
We selected the current loop crossover frequency after compensation
f
_{ci}
= 0.1˗0.2
f
_{s}
in these transfer functions. We then placed the zero
f
_{z1}
of
G
_{i}
(
s
) at the output resonance frequency
f
_{0}
for damping and pole
f
_{p1}
=
f
_{s}
/10 for switching ripple elimination.
At this point, the current loop gain magnitude at
f
_{ci}
before compensation is ｜
T
_{i_o1}
(
j
·2
πf_{c}
)｜
dB
, which indicates that the compensation gain should be as follows:
which ensures that the current loop gain magnitude at
f
_{ci}
after compensation is zero.
Furthermore,
Considering the component tolerances, the resistors and capacitors whose values are near the calculated ones are selected and then reexamined by the Bode plots, as shown in
Fig. 8
. The current loop crossover frequency is
f
_{ci}
= 2.9 kHz, and the phase margin is 40°, which means that the current loop has a suitable stable and dynamic performance.
Loop gains after compensation.
After the current loop is closed, the inner loop is used as a power stage as follows:
The outer loop gain before compensation is as follows:
The outer loop controller and outer loop gain can be designed similar to the previously mentioned gains. The outer loop gain after compensation is described as shown in Eq. (21), and more information is provided in
Fig. 8
.
(current loop cross over frequency
f
_{ci}
= 2.9 kHz, PM = 40°,
R
_{4}
=
R
_{5}
= 10 kΩ,
R
_{6}
= 6.2 kΩ,
C
_{3}
= 47 nF,
C
_{4}
= 13 nF; outerloop crossover frequency
f
_{cv}
= 390 Hz, PM = 80°,
R
_{1}
=
R
_{2}
= 10 kΩ,
R
_{3}
= 30 kΩ,
C
_{1}
= 9.1 nF,
C
_{2}
= 2.7 nF)
V. EXPERIMENTAL VERIFICATION
 A. Hardware Description
A 1.2 kW hardware prototype for a marine battery charger was designed, fabricated, and tested to verify the proposed converter and currentsharing strategy. The final charger product started its type test. Detailed specifications and parameters are shown in
Fig. 9
and
Table I
. A singlechip Atmega64 controller provides the voltage and current reference for battery charge management, while a TI UCC3895 IC controls the FB circuit.
Prototype photograph. (a) 3D view of the virtual prototype. (b) Interior of the actual prototype.
OPERATION CONDITIONS AND CIRCUIT PARAMETERS
OPERATION CONDITIONS AND CIRCUIT PARAMETERS
 B. Experimental Key Waveforms
Fig. 10
provides the detailed experimental results of the topology shown in
Fig. 2
. The fast reset of the primary current is observed in
Fig. 10
(a), which implies that the circulating current is eliminated, thus helping in efficiency improvement.
Experimental results.
Fig. 10
(b) illustrates the FB primary middlepoint voltage and transformer primary voltage. The difference between the voltage drops in the blocking capacitor
C
_{B}
is shown in
Fig. 10
(c).
Fig. 10
(d) shows the ZCS operation of the IGBTs in the lagging leg with load current adaptability. Notably, the device current drops to zero at the lightload condition before the gate signal is turned off. Thus, the ZCS is achieved at this point, although the ZVS condition of the lagging leg is still not obtained as shown in
Fig. 10
(e).
Fig. 10
(f) shows the test using the simple phaseshift modulation method.
 C. Experimental Data and Discussion
Fig. 11
further provides the efficiency curve of the prototype. The expected high efficiency is guaranteed because of the topology with intrinsic soft switching and eliminated circulating current features. The maximum efficiency is approximately 93% under 540 Vdc input and 28 Vdc output condition, where auxiliary power and fan losses are also included.
Efficiency curve.
The decrease in efficiency for currents of 30 A up to 40 A is caused by conduction losses, especially the output rectifying diodes. The impact of conduction losses is highly significant in converters with low output voltage and high output current, such as in our case. In addition, no energy recovery circuits are added in the converter to clamp rectify diode voltage spikes and to achieve forward and reverse recovery current optimization.
Nevertheless, the efficiency is relatively high, unlike the ZVS FB converter in
Fig. 11
. The only differences between these converters are that the IGBTs without body diodes in the lagging leg use the same devices with body diodes in the leading leg. Highoutput voltage and lowoutput current will be higher, especially in the case where the synchronous rectification technique or lossless energy recovery clamp circuits are introduced
[30]
.
Fig. 12
shows the hotspot temperature curves of the prototype devices. Thermal balance is achieved after 30 min of work. The maximum temperature increase is approximately 20 ℃ for primary switches and secondary diodes. The hottest component is the output choke, which is insensitive to the heat.
Prototype devices with hotspot temperature curves (ambient temperature: 25 ℃).
Table II
further summarizes the current distribution for a parallel connection of two prototype modules. The currentsharing accuracy is in the 1% to 6% range in the entire load range with the currentsharing bus connected. Even at a lightload condition, the current distribution is insensitive to the noise. The error is mainly caused by the mismatching in circuit power stages because the sharing bus provides the same DC current reference. The implementation of the true
N
+ 1 redundant system can then be conducted
[25]
.
DATA OF PARALLEL CURRENT SHARING
DATA OF PARALLEL CURRENT SHARING
Table III
further provides a comparative study of the possible prototype candidates. The proposed topology exhibits a suitable balance between performance and cost for industry application with the least components and devices in different possibilities.
EVALUATION CONDITIONS AND RESULTS
EVALUATION CONDITIONS AND RESULTS
VI. CONCLUSIONS
A novel inherent ZVZCS PS FB converter is proposed in this paper. The operation principles, specific design considerations, and experimental results are presented. The distinctive features of the proposed topology are summarized as follows.

1) The ZVS transition of the leading leg is supported by the secondary side filter inductanceLfand the transformer leakage inductanceLlk. Thus, the ZVS range of the leading leg is relatively wide.

2) The ZCS transition of the lagging leg and minimized circulating current can be achieved in the total line input and output load range through the use of IGBTs with the RB feature. Therefore, the turnoff and conduction losses of the IGBTs are significantly lowered.

3) The NPTIGBT without a body diode can be a lowcost solution to replace RB IGBT for mass production. The optimization of switching performance of the RB IGBT is no longer a key issue that makes RB IGBT unacceptable in real applications. Common IGBT drivers are sufficient.

4) The low topologycomplexity FB converter without any auxiliary circuit can be easily controlled with a simple PS PWM control strategy. A simple, lowcost, and robust democratic currentsharing circuit is introduced and verified in this study, which offers redundancy implementation, expandability of output power, and ease of maintenance features. This converter is an attractive alternative for compact and costeffective applications with highvoltage input.
Acknowledgements
This work was supported in part by the National Natural Science Foundation of China (51207023) and the 2012 Special Fund for the Development of Small and MediumSized Enterprises of Zhenjiang, China.
BIO
Jianhua Wang was born in Nantong, China. He received his B.S. and Ph.D. in Electrical Engineering from Nanjing University of Aeronautics and Astronautics, Nanjing, China, in 2004 and 2010 respectively. He joined the faculty of the School of Electrical Engineering in Southeast University, Nanjing, China, in 2010, where he is currently a lecturer. He has published more than 30 technical papers and is the holder of two China patents. His main research interests are general power electronic circuit topologies, modeling, control, and power electronics system stability, solidstate power electronics transformer, and highperformance power conversions for renewable energy, ship, vehicle, and aerospace applications.
Baojian Ji was born in Yancheng, China. He received his B.S. in Automation Engineering from Nanjing Normal University, Nanjing, China, M.S. in Electrical Engineering from the Nanjing University of Aeronautics and Astronautics, Nanjing, and Ph.D. in Electrical Engineering from the Southeast University, Nanjing in 2002, 2007, and 2012, respectively. He joined Nanjing University of Technology as a lecturer in 2007, where he is currently the Head of the Department of Electrical Engineering. He has published more than 20 technical papers. His research interest includes the digital control technique and development of a gridtied inverter for renewable energy applications.
Hongbo Wang was born in Yichun, China. She received her B.S. in Electrical Engineering from Jiangsu University, Zhenjiang, China, in 1992. She is now the vicechief engineer of the System Integration Department of SaierNico Electric & Automation LTD., Zhenjiang, China. Her research interests are marine automation and switchboard systems.
Naifu Chen was born in Yancheng, China. He received his B.S. and M.S. in Electrical Engineering from Nanjing University of Technology, Nanjing, China in 2007 and 2014 respectively. He joined Shanghai Acrel LTD., Shanghai, China, in 2014, where he is currently an R&D Engineer. His research interests are battery chargers and PV inverters.
Jun You was born in Nanjing, China. He received his B.S. in Automation from Nanjing University of Aeronautics and Astronautics, Nanjing, China in 1998. He received his M.S. in Power Electronics and Electrical Drives and Ph.D. in Electrical Engineering from Southeast University, Nanjing, China in 2001 and 2012 respectively. He has been with the School of Electrical Engineering, Southeast University since 2001, where he is currently an associate professor. He has also been the Deputy Director of the Suzhou Key Laboratory of Electrical Equipment and Automation of Research Institute of Southeast University in Suzhou, China, since 2010. His main research interests include power electronics, gridconnected renewable energy systems, and power quality monitoring.
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