For lowpower highfrequency LED driver applications in small form factor mobile products, a highefficiency boundary conduction mode tappedinductor boost converter is proposed. In the proposed converter, the switch and the diode achieve softswitching, the diode reverserecovery is alleviated, and the switching frequency is very insensitive to output voltage variations. The circuit is quantitatively characterized, and the design guidelines are presented. Experimental results from an LED backlight driver prototype for a 14 inch notebook computer are also presented.
I. INTRODUCTION
Liquid crystal displays (LCD) are currently in a very wide use as flat panel display devices, whose diagonal length ranges from subinch to more than 100 inches. Since LCDs cannot generate light by themselves, they need an illumination unit to make an image visible. This illumination unit is often called a backlight unit, and light emitting diodes (LED) are recently achieving a dominant position as this light source since they have many excellent features such as high energy efficiency, high scalability, long lifetime, and ecofriendliness when compared to conventional coldcathode fluorescent lamps
[1]

[5]
.
In general, a smaller backlight unit tends to have a smaller number of LEDs, as shown in
Fig. 1
(a). However, this is not always true since design requirements are becoming more stringent and require a narrower bezel. Due to the pointsource nature of LEDs, narrowbezel products require a significantly larger number of LEDs than widebezel products to maintain good brightness uniformity for the same display area. This is shown in
Fig. 1
(b). Even a smallsized backlight unit whose diagonal length is only a few inches can be easily found to have as many as several tens of LEDs. In addition, the voltage required to drive them often exceeds 50V.
The conventional boost converter shown in
Fig. 2
is widely being used as, or as a part of, constant current LED drivers for the backlight units where the required output voltage is higher than the available input voltages to the circuit. It is preferable for this converter to operate in continuous conduction mode (CCM) to minimize the conduction loss caused by a high input current ripple
[6]
,
[7]
. However, in low power applications such as the LED backlight drivers for small form factor mobile products, the conduction loss is not that critical and often considered to be insignificant since the input current is relatively small. In such cases, operation in boundary conduction mode (BCM), which is also known as critical conduction mode (CRM), finds a good opportunity since it features no diode reverserecovery problem, no right halfplane zero, and no need for slope compensation. Furthermore, the switch and the diode can respectively achieve zero voltage switching (ZVS) and zero current switching (ZCS) through the resonance of the boost inductor and the output capacitance of switching devices if appropriate operating conditions are provided
[6]

[15]
.
The biggest problem with the BCM boost converter for LED driver applications is that even a small variation in the output voltage causes a significant change in the switching frequency. Unfortunately, the voltage across the LED is given as a ranged value due to its dependencies on the temperature, the material used, the manufacturer, the production lot, and so on
[3]

[5]
. To make matters worse, the LED voltage can change even beyond its specified normal range due to changes in the material properties through longterm aging.
C_{o}
nsequently, the BCM boost LED driver needs to manage wide variations in the switching frequency, which makes it difficult to optimize the circuit from the standpoint of efficiency, transient response, ripple content, electromagnetic interference (EMI), and so on. This, in spite of its many attractive features, restricts the use of the BCM boost converter as an LED driver.
Typical LED backlight unit.
Conventional boost LED driver.
As an alternative to the conventional BCM boost converter, a BCM tappedinductor boost (TIB) converter for LED driver applications is proposed in this paper. When compared to the conventional BCM boost converter, the proposed converter shows a switching frequency which is very insensitive to changes in the output voltage
[15]
. It also shows a higher stepup ratio and a lower switch voltage stress than its conventional counterpart
[16]

[18]
. Moreover, the proposed converter provides ZVS of the switch and ZCS of the diode, which allows for a higherfrequency operation with a more compact circuit implementation. Therefore, the proposed BCM TIB converter is considered to be a good candidate for constant current LED driver applications especially for small form factor mobile products. In this paper, the steadystate and softswitching operations of the proposed converter are analyzed, various voltage and current stresses are quantitatively characterized, and a design guideline optimizing the tappedinductor loss is presented. Experimental results from a prototype driver circuit for a 14 inch LED backlight for a notebook computer are also presented.
II. ANALYSIS OF THE BCM TIB LED DRIVER
A schematic diagram and the key operating waveforms of the proposed BCM TIB converter are shown in
Fig. 3
and
Fig. 4
, respectively. Switch
M
_{1}
operates at a frequency of
f
=1/
T
and a duty ratio of
D
. The tappedinductor is modeled as an ideal transformer with a turns ratio of
N
=
N_{s}
/
N_{p}
, a magnetizing inductance
L_{m}
, which is referred to the primary side, and a leakage inductance
L_{k}
, which is referred to the primary side. The value of
L_{m}
is assumed to be much larger than that of
L_{k}
. The input voltage
V_{i}
, the output voltage
V_{o}
, and the output current
I_{o}
are assumed to be ripplefree and constant. For convenience in developing equations, 1+
N
is denoted by
k
.
Tappedinductor boost (TIB) topology.
Key operating waveforms.
 A. Steadystate Analysis
The topological stages according to the switch state are shown in
Fig. 5
. Initially, switch
M
_{1}
is assumed to be in the offstate. It is turned on as soon as the decreasing magnetizing current
i_{m}
reached zero at
t
=0.
Onmode [0~DT]:
This mode begins when switch
M
_{1}
turns on. Since the output diode
D
_{1}
is reversebiased, there is no induced current in the coupled windings, and all of the input current, denoted by
i_{i}
, flows through
L_{k}
and
L_{m}
. From the assumption that
L_{m}
≫
L_{k}
, the leakage inductance
L_{k}
in the input current loop can be neglected, and the voltage across
L_{m}
is equal to
V_{i}
. The magnetizing current
i_{m}
, which is equal to
i_{i}
, increases linearly from zero to
I_{pk}
.
The voltage stress on
D
_{1}
, shown by
V_{ka}
in
Fig. 4
, is equal to the sum of
V_{o}
and
V_{i}
reflected to the secondary side. It can be normalized with respect to
V_{i}
as:
Offmode [DT~T]:
This mode begins when switch
M
_{1}
turns off at
t
=
DT
. Due to the leakage inductance
L_{k}
, the switch current cannot instantaneously fall to zero the moment
M
_{1}
turns off. Therefore, a high voltage spike can be developed at the switch drain, as illustrated in
Fig. 4
. To alleviate this, the tappedinductor windings need to be tightly coupled, or a snubber circuit should be considered.
After a brief transient due to
L_{k}
, switch
M
_{1}
enters the offstate, and diode
D
_{1}
enters the onstate, as shown in
Fig. 5
(b). The input current
i_{i}
is equal to the secondary current
i_{s}
, and the magnetizing current
i_{m}
is equal to sum of the input current
i_{i}
and the primary induced current of
N
·
i_{s}
, resulting in
i_{m}
=
k
·
i_{i}
. The input current
i_{i}
decreases linearly from
I_{pk}
/
k
to zero during this mode as the magnetizing current
i_{m}
decreases linearly from
I_{pk}
to zero. Since the equivalent inductance of the tappedinductor in this stage is
L_{k}
+
k
^{2}
L_{m}
[16]
, the leakage inductance
L_{k}
, which is assumed to be much smaller than
k
^{2}
L_{m}
, can be safely neglected.
The winding voltages are proportional to their number of turns. Therefore, the voltage across
L_{m}
can be obtained to be (
V_{i}

V_{o}
)/
k
. Subtracting this voltage from
V_{i}
gives the voltage stress on
M
_{1}
, shown by
V_{ds}
in
Fig. 4
, which can be normalized with respect to
V_{i}
as:
This mode ends when the input current reaches zero and the tappedinductor is demagnetized.
Neglecting the effect of
L_{k}
, whose contribution to the stepup ratio is very small, the stepup ratio of the BCM TIB converter and the inverse function can be obtained by applying the voltsecond balance to
L_{m}
for one switching cycle as:
which is identical to the CCM TIB converter.
Topological stages.
Since the average current in the output capacitor
C_{o}
is zero in the steadystate, the relationship between
I_{pk}
and
I_{o}
can be obtained by averaging
i_{s}
for one switching cycle to equate with
I_{o}
as:
where
D
is eliminated using (3).
The curves of
D
,
I_{pk}
/
I_{o}
,
V_{ds}
/
V_{i}
, and
V_{ka}
/
V_{i}
, given by (3), (4), (2), and (1) respectively, are plotted in
Fig. 6
as functions of
V_{o}
/
V_{i}
for different values of
N
. Since the proposed converter is supposed to supply a constant current to an LED string whose forward voltage is given as a ranged value, it is reasonable to study the characteristics as functions of
V_{o}
/
V_{i}
rather than as functions of the duty ratio.
Fig. 6
and the corresponding equations show that decrements of
D
and
V_{ds}
/
V_{i}
, over an increment of
N
at a given
V_{o}
/
V_{i}
, decrease as
N
increases. Therefore,
D
and
V_{ds}
/
V_{i}
converge to 0 and 1 respectively when
N
keeps increasing. On the other hand, increments of
I_{pk}
/
I_{o}
and
V_{ka}
/
V_{i}
, over an increment of
N
at a given
V_{o}
/
V_{i}
, do not vanish as
N
increases. Therefore,
I_{pk}
/
I_{o}
and
V_{ka}
/
V_{i}
consistently increase as
N
increases. This means that the value of
N
should not be too large. If it is, the disadvantages from the increased
I_{pk}
and
V_{ka}
overtake the advantages from the decreased
D
and
V_{ds}
.
The switching frequency
f
of the BCM TIB converter can be obtained by equating
I_{pk}
, given by (4), with the increase in
i_{m}
during Onmode as:
where
D
is eliminated using (3).
Tendencies of key values to V_{o}/V_{i} variation.
For the best visualization of (5), the operating conditions are exemplified as
V_{i}
=14V,
V_{o}
=55V,
I_{o}
=44mA, and
f
=220kHz. The values of
L_{m}
satisfying these conditions are determined using (5) for different values of
N
. Then, the switching frequencies of the proposed converter for different pairs of
N
and
L_{m}
are calculated from (5) as functions of
V_{o}
. The results are plotted in
Fig. 7
. The curve for the conventional BCM boost converter, shown by a dashdot line, is obtained from (5) where
N
=0. It is obvious that the conventional BCM boost converter has a steep negative slope at the operating point, whereas the proposed BCM TIB converter has a much more gradual slope with a positivegoing tendency as
N
increases. This characteristic makes the proposed converter very attractive for LED driver applications since the switching frequency dependent aspects such as the efficiency, the ripple content, the EMI, and the transient response can be made very stable.
By differentiating (5) with respect to
V_{o}
and equating the result with zero, the stationary point of (5), where the frequency change due to a small perturbation in
V_{o}
is zero, can be found at:
Tendency of switching frequency to V_{o} variation.
 B. Softswitching analysis
Due to the BCM operation, the output diode
D
_{1}
in the Offmode is turned off not by a change in the voltage across it but by the current gradually decreasing to zero, as shown in
Fig. 4
. As a result, the ZCS operation of
D
_{1}
is achieved, and the reverserecovery current of
D
_{1}
is alleviated.
The moment
D
_{1}
turns off, all of the currents
i_{i}
,
i_{m}
, and
i_{s}
reach zero, and the output capacitance of the switching devices begins to resonate with the magnetizing inductance. The resonant elements are identified by dashed rectangles in
Fig. 8
, where the leakage inductance
L_{k}
is again neglected since the resonant behavior is governed by
L_{m}
which is much larger than
L_{k}
.
The resonant waveforms of
v_{ds}
,
i_{i}
, and
v_{ka}
are illustrated in
Fig. 9
. The input current
i_{i}
is assumed to reach zero at
t
=
t
_{zc1}
, and
M
_{1}
is assumed to turn on at
t
=
t
_{zc2}
, the moment
i_{i}
crosses zero to change its direction. The solid line shows a case where the ZVS of the switch can be provided since the resonant amplitude of
v_{ds}
is large enough to touch zero at
t
=
t_{zv}
. In this case, the voltage
v_{ds}
cannot decrease further below zero after
t
=
t_{zv}
since the body diode of
M
_{1}
turns on. If
M
_{1}
turns on at any time between
t
=
t_{zv}
and
t
=
t
_{zc2}
while the voltage
v_{ds}
is clamped at zero, the ZVS of
M
_{1}
can be achieved. The dashed lines in
Fig. 9
illustrate the opposite case where the voltage
v_{ds}
cannot reach zero, and the ZVS of
M
_{1}
cannot be achieved.
To derive the ZVS condition for
M
_{1}
, a set of differential equations are set up for the resonant elements as:
Substituting (8) and (9) into (7) results in:
Substituting
v_{m}
into (10) with
V_{i}

v_{ds}
results in a second order differential equation for
v_{ds}
as:
where
v_{ds}
(0)=(
V_{o}
+
NV_{i}
)/
k
. By solving (11), the equation for
v_{ds}
can be obtained as:
Resonant circuit after tappedinductor demagnetization.
Waveforms after tappedinductor demagnetization.
where
ω
={
L_{m}
(
C_{ds}
+
k
^{2}
C_{ka}
)}
^{1/2}
. Since
v_{ka}
is equal to
V_{o}
+
Nv_{m}

v_{ds}
and
v_{m}
is equal to
V_{i}

v_{ds}
, the equation for
v_{ka}
can be obtained as:
To provide the ZVS of
M
_{1}
, the minimum value of (12) should be negative, which results in the ZVS condition for
M
_{1}
as:
where
V_{o}
/
V_{i}
is eliminated by using (3).
It should be noted that
v_{ka}
stays at
V_{o}
+
NV_{i}
while
v_{ds}
is clamped at zero. Since this voltage is equal to the diode turn off voltage given by (1), a fast
dv_{ka}
/
dt
associated problems such as the EMI due to the spiky charging current of
C_{ka}
at the moment of the switch turnon is also mitigated if (14) is satisfied.
 C. Current stress analysis
Since the switching losses in the proposed converter are very small due to the softswitching operation, the conduction loss due to the parasitic resistance in the circuit is the next one to be considered. Neglecting the effect of
L_{k}
, the major RMS current stresses normalized with respect to
I_{o}
can be calculated from the waveforms in
Fig. 4
as:
By eliminating
D
in (15)(18) using (3), the curves of
I
_{i,RMS}
/
I_{o}
,
I
_{d,RMS}
/
I_{o}
,
I
_{s,RMS}
/
I_{o}
, and
I
_{c,RMS}
/
I_{o}
are plotted in
Fig. 10
as functions of
V_{o}
/
V_{i}
for different values of
N
. Since the duty ratio
D
given by (3) approaches zero as
N
increases at a given
V_{o}
/
V_{i}
, both
I
_{s,RMS}
/
I_{o}
and
I
_{c,RMS}
/
I_{o}
finally converge to constant values defined by
D
=0 when
N
keeps increasing. On the other hand, the increments of
I
_{i,RMS}
/
I_{o}
and
I
_{d,RMS}
/
I_{o}
over an increment of
N
at a given
V_{o}
/
V_{i}
do not disappear as
N
increases. Therefore,
I
_{i,RMS}
/
I_{o}
and
I
_{d,RMS}
/
I_{o}
consistently increase as
N
increases. This means that, if the value of
N
is too large, the disadvantages from the increased
I
_{i,RMS}
and
I
_{d,RMS}
overtake the advantages from the decreased
I
_{s,RMS}
and
I
_{c,RMS}
.
Trend of normalized RMS currents to V_{o}/V_{i} variation.
III. DESIGN CONSIDERATION
It is often the case that presenting a design guideline with a practical design example is more convenient and illustrative. The requirements for the prototype BCM TIB LED driver are assumed as
V_{i}
=14V,
V_{o}
=55V±15%, and
I_{o}
=44mA for the 14 inch LED backlight of a notebook computer. The parameters to be given and to be determined are summarized in
Table I
.
First, the tappedinductor turns ratio can be determined by using the conditions for the switching frequency stability and the ZVS of the switch. The value of
N
for the best frequency stability at the typical output voltage of 55V can be determined from (6) to be 1.93. On the other hand, to ensure the ZVS over the entire output voltage range in an actual circuit, an extra 10% margin is applied at the minimum output voltage, resulting in
N
<1.01 from (14). It should be noted that both of these conditions cannot be satisfied at the same time. Since efficiency is more critical in mobile products, the condition for the ZVS takes a higher priority. As a result, the switching frequency stability needs to be sacrificed to some extent. Therefore, the value of
N
is finally chosen to be 1.
DESIGN PARAMETERS
It should be noted that the switching frequency range of the proposed BCM TIB LED driver to the entire output voltage range is estimated by using (5) to be 4.2/+3.0% of a typical value, while that of the conventional BCM boost LED driver is estimated to be 9.2/+11%. This shows that the switching frequency stability is still significantly better in the proposed converter, even though it is somewhat sacrificed in choosing the value of
N
.
It is generally desired that the switching frequency be as high as possible to minimize the values and sizes of the reactive components. However, this is limited by a decrease in efficiency due to increases in the magnetic core loss, the switch turnoff loss, and the switch gate drive loss. Moreover, a higher frequency of operation also causes higher levels of various nonideal sideeffects such as the skin effect, proximity effects, and EMI, which are very difficult to predict quantitatively. Therefore, the switching frequency is frequently chosen on an empirical base rather than from an optimization process. In this example, the typical switching frequency at the typical output voltage condition of
V_{o}
=55V is chosen to be
f
=220kHz in consideration of the aforementioned sideeffects.
The magnetizing inductance of the tappedinductor can now be obtained from (5) as
L_{m}
=87
μ
H at the typical output voltage of 55V. The number of turns of the primary winding
N_{p}
and the air gap length
l_{g}
of the tappedinductor can be obtained from the following two equations:
where
μ
_{0}
is the permeability of vacuum equal to 4π×10
^{7}
,
A_{c}
is the core crosssectional area at the air gap which is given by the core manufacturer as
A_{c}
=7.1mm
^{2}
, and
B_{pk}
is the peak limit of the core flux density. Considering the large flux swing due to the BCM operation, it is a good practice to make the value of
B_{pk}
equal to half the saturation flux density. Assuming a core made of MGB1 ferrite from TAK Technology whose saturation flux density is 0.32T, the value of
B_{pk}
is chosen to be 0.16T. Under the maximum output voltage condition, (4), (19), and (20) can be solved to give
N_{p}
=37 and
l_{g}
=0.14mm. According to the tappedinductor manufacturer’s standards, the value of
l_{g}
is finally adjusted to the closest large standard value of 0.15mm, and the value of
N_{p}
is adjusted to 38 to maintain the value of
L_{m}
. The number of turns of the secondary winding
N_{s}
can be obtained from
N_{s}
=
N
·
N_{p}
as 38.
The wire loss in the tappedinductor can be obtained as:
where the definitions and values of
ρ
,
MLT_{p}
,
MLT_{s}
,
A_{p}
, and
A_{s}
are listed in
Table I
. By using the fill factor constraint defined by:
where
K_{u}
is the winding fill factor and
A_{w}
is the core window area,
A_{s}
in (21) can be eliminated to give:
By differentiating (23) with respect to
A_{p}
, the stationary point where the minimum of (23) occurs can be found by solving:
together with (3), (15) and (17) to give:
By substituting the given and determined values listed in
Table I
into (25), the value of
A_{p}
minimizing the wire loss in the tappedinductor is obtained as
A_{p}
=0.082mm
^{2}
. The value of
A_{s}
is obtained from (22) as
A_{s}
=0.029mm
^{2}
. The diameters of the primary and secondary windings are calculated to be 0.32mm and 0.19mm respectively. However, they are adjusted to 0.3mm and 0.2mm to allow for the use of standard wires.
Finally, the core loss of the tappedinductor can be estimated from the core manufacturer’s data sheet by using the switching frequency
f
and the peak flux density
B_{pk}
. In the case of too much core loss, increasing the air gap length
l_{g}
to reduce
B_{pk}
is worth trying as a quick remedy. However, it should be noted that this requires more winding turns to maintain the original inductance value, resulting in an increase in the wire loss. If both the core loss and the wire loss are unacceptably high, a different core with bigger size should be considered.
IV. EXPERIMENTAL RESULTS
A 44mA, 14V/55V±15% prototype of the proposed BCM TIB LED backlight driver for a 14 inch notebook computer has been built as described in the previous chapter. The control circuit has been implemented with the peak current mode control scheme. The switch turns off when the switch current exceeds the output current error signal generated by an error amplifier and it turns back on when the switch voltage becomes zero to achieve ZVS. The switch current is sensed by a resistor at the MOSFET source, and the switch voltage is sensed by a resistor divider at the MOSFET drain. A 60Vrated IRFL014 with
R
_{DS(ON)}
=0.2Ω is chosen for the switch. A 100Vrated 1N4934 is chosen for the diode. Since the tappedinductor turns ratio is 1, a bifilar winding is used to minimize the leakage inductance.
Fig. 11
shows a photo of the implemented setup. The LEDs are mounted on a flexible printed circuit (FPC). A small PCB for interfacing the FPC and wires are used in the setup.
Fig. 12
shows the key experimental waveforms. Referring to
Fig. 9
, the moment the input current
i_{i}
reaches zero is denoted by
t
=
t
_{zc1}
. The switch voltage
v_{ds}
and the input current
i_{i}
start resonating at
t
=
t
_{zc1}
, and the voltage
v_{ds}
reaches zero at
t
=
t_{zv}
. The gate turns on at
t
=
t_{zv}
to achieve ZVS. Due to the small tappedinductor leakage inductance, whose measured value is 0.2
μ
H, the turnoff spike in the
v_{ds}
waveform is small enough to avoid the use of a snubber circuit.
Implemented setup.
Experimental waveforms.
The switching frequency is measured to be 200kHz, which is slightly lower than the designed value of 220kHz. This is because the gate turnon delay introduced to provide ZVS, which is not considered in deriving (5), reduces the switching frequency
[13]
. In addition to the gate turnon delay, both the switch ontime and the switch offtime increase to maintain the averaged input current. This results in a small additional decrease in the switching frequency.
For comparison purposes, a conventional BCM boost LED driver has been made by replacing the tappedinductor in the proposed BCM TIB LED driver prototype with an inductor using the same core. The inductor value is chosen to be 137
μ
H to maintain the typical operating point shown in
Fig. 7
. The switch is also replaced with a 100Vrated switch since the voltage stress in the conventional driver at the maximum output voltage exceeds the maximum rating of the IRFL014. A FQT7N10L with
R
_{DS(ON)}
=0.35Ω is chosen since the
Q_{g}
×
R
_{DS(ON)}
figure of merit (FOM) of 2.1 is very close to that of the IRFL014 at 2.2.
Fig. 13
shows the efficiencies of both of the LED drivers according to the output current
I_{o}
. At the typical output current of 44mA, the proposed BCM TIB LED driver shows an efficiency of 96.4%, and the BCM boost LED driver shows an efficiency of 95.9%. When the output current is low, the switch turnoff loss becomes dominant with a high switching frequency. When the output current is high, the conduction loss becomes dominant with a high input current. These are reflected in
Fig. 13
. Since the low switch voltage stress of the proposed LED driver allows for the use of switching devices with a low voltage rating and a low
R
_{DS(ON)}
, the proposed LED driver shows higher efficiency than the BCM boost LED driver.
Measured efficiencies.
Fig. 14
shows the ideal switching frequencies of both of the LED drivers predicted by (5) and the experimental results. It confirms that the switching frequency of the proposed LED driver is much more stable with respect to variations in the output voltage. It also shows that the gap between the ideal and the experimental curves are larger in the BCM boost LED driver. This is because the largervalued inductance of the BCM boost LED driver requires a longer gate turnon delay to achieve ZVS.
Ideal and experimental switching frequencies.
V. CONCLUSIONS
In lowpower stepup LED driver applications where the input current is so small that the associated conduction loss is not very critical, a BCM boost converter is a good candidate since the softswitching feature allows for an increase in the switching frequency to achieve a compact circuit design. However, the conventional BCM boost topology often fails in such applications since variations in the LED voltage causes a significant change in the switching frequency.
To overcome this problem, a BCM TIB LED driver whose switching frequency is very insensitive to changes in the output voltage is proposed. It also features softswitching of both the switch and the diode, no diode reverserecovery problem, a high stepup ratio, and a low switch voltage stress. Therefore, the proposed BCM TIB LED driver is considered to be very promising in small form factor mobile product applications.
BIO
Jeongil Kang received his B.S., M.S., and Ph.D. degrees in Electrical Engineering from the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea, in 1995, 1997, and 2002, respectively. In 2002, he joined Samsung Electronics Co., Ltd., Suwon, Korea, and is currently working on the design and control of semiconductor light source drivers for various display and lighting devices as a Principal Engineer in the Visual Display Research & Development Office. His current research interests include the development, modeling, and control of power converter topologies. He is a Member of the Korean Institute of Power Electronics (KIPE).
SangKyoo Han received his M.S. and Ph.D. degrees in Electrical Engineering and Computer Science from the Korea Advanced Institute of Science and Technology (KAIST), Daejeon, Korea, in 2001 and 2005, respectively. For the next six months, he was a PostDoctoral Fellow at KAIST where he developed digital display power circuits and preformed several research activities. Since 2005, he has been with the Department of Electrical Engineering, Kookmin University, Seoul, Korea, as an Associate Professor. He has also worked for the Samsung Power Electronics Center (SPEC) and the Samsung Network Power Center (SNPC) as a Research Fellow. His primary areas of research interests include power converter topologies, LED drivers, renewable energy systems, and battery chargers for electric vehicles. Dr. Han is a Member of the Korean Institute of Power Electronics (KIPE).
Jonghee Han is an Executive Vice President of the Visual Display Business for Samsung Electronics. In this role, he leads the Visual Display Research & Development Office. Since joining Samsung in 1988, Mr. Han has played a significant role in the success of Samsung’s Visual Display business, helping it solidify its top position within the consumer electronics industry. One of his many successes includes the launch of the awardwinning ‘Bordeaux’ TV series, which became one of the world’s bestselling LCD TVs in 2006. During his tenure, Mr. Han has accomplished a number of “firsts” in the TV industry, serving at the forefront of some of the most noteworthy TV advancements. For example, he was an integral force in the development of the world’s first 3D LED TV in 2010 and first Smart TV in 2011, among numerous other developments. More recently, he played a critical role in the creation of both the world’s first Curved OLED TV and Curved UHD TV. In particular, his research specialties include image processing, audio signal processing, analog/digital mixed systems, display devices, and power electronics for both innovative displays and audio products. Along with contributing his vast knowledge to Samsung’s Visual Display Business, he is also a Member of the Korean Institute of Power Electronics (KIPE). Mr. Han holds a bachelor’s degree in electrical engineering from Inha University, Incheon, Korea.
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