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A 3 kW Bidirectional DC-DC Converter for Electric Vehicles
A 3 kW Bidirectional DC-DC Converter for Electric Vehicles
Journal of Electrical Engineering and Technology. 2016. Jul, 11(4): 860-868
Copyright © 2016, The Korean Institute of Electrical Engineers
This is an Open-Access article distributed under the terms of the Creative Commons Attribution Non-Commercial License (http://creativecommons.org/licenses/by-nc/3.0/) which permits unrestricted non-commercial use, distribution, and reproduction in any medium, provided the original work is properly cited.
  • Received : June 09, 2015
  • Accepted : October 29, 2015
  • Published : July 01, 2016
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About the Authors
Arsalan, Ansari
Dept. of Electronic Systems Engineering, Hanyang University, Korea.
Puyang, Cheng
Dept. of Electronic Systems Engineering, Hanyang University, Korea.
Hee-Jun, Kim
Corresponding Author: Dept. of Electronic Systems Engineering, Hanyang University, Korea.

Abstract
A bidirectional DC-DC converter (BDC) is an indispensable electrical unit for the electric vehicles (EVs). High efficiency, high power density, isolation, light weight and reliability are all essential requirements for BDC. In this paper, a 3 kW BDC for the battery charger of EVs is proposed. The proposed converter consists of a half-bridge structure on the primary side and an isolation transformer and a synchronous rectifier structure on the secondary side. With this topology, minimum number of switching devices are required for bidirectional power flow between the two dc buses of EVs. The easy implementation of the synchronous rectification gives advantages in terms of efficiency, cost and flexibility. The proposed BDC achieves high efficiency when operating in both modes (step-up and step-down). A 3 kW prototype is implemented to verify theoretical analysis and the performance of the proposed converter.
Keywords
1. Introduction
With the global energy crisis the conventional vehicles (internal combustion engines) face the increasingly serious problems of energy. In contrast, the EVs especially battery electric vehicles (BEVs) depend on variety of options for its driving power. BEVs offer the advantages of safety, silent operation and no emissions when powered by renewable energy sources such as wind or solar which are virtually emission free [1] . These vehicles can also make efficient use of energy by storing energy recovered during braking or deceleration cycle in the batteries. The storage or charging process of the battery is achieved by a BDC, which is the key block in EV energy system to link high voltage (HV) dc bus and low voltage (LV) dc bus as shown in Fig. 1 . This BDC should have high power density and high efficiency to meet the desired goals for EV’s battery charger. When the EV is parked, the battery can be charged by the household utility outlet from the grid through the BDC. For the other case when the EV is in the driving state, the BDC provides the electrical power from LV battery to the motor through DC-AC inverter and also DC loads in the EV.
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The energy system of EV
BDCs are broadly classified into isolated and non-isolated types. The conventional non-isolated buck/boost BDC cannot operate in the wide voltage conversion range [2] . The isolated BDC are preferred for EVs due to the advantages of high voltage conversion ratio and safety. Many different types of isolated BDCs [3 - 6] have been proposed due to these advantages, some full-bridge BDCs [7 - 10] have also been proposed in recent years. However, full-bridge converters have the disadvantage of high voltage ripples if not employing an extra voltage clamping circuit [11] . By contrast, half-bridge converters [12 - 14] have a simple structure and a better anti-imbalance ability in the transformer. In some topologies of halfbridge converters, MOSFET body diodes are applied for synchronous rectification in both buck/boost modes [15] , but high conduction losses result in low efficiency, thus limiting the use of these converters to only low power applications.
This paper describes the development of a 3 kW BDC for EVs. The converter consists of a half-bridge topology, an isolation transformer and a synchronous rectifier. The isolation transformer provides the advantages of wide conversion range and safety, and the easy implementation of synchronous rectification offers the benefits in terms of efficiency, cost and flexibility. However, this structure has been mostly proposed for less than 1.5 kW application [3 , 11 - 13 , 16 - 17] , so the operation of more than 3 kW in both step-up and step-down modes has the practical significance for the EV battery charger products.
2. Topology Configuration and Operational Principles
- 2.1 Topology configuration
Fig. 2 shows the circuit diagram of the proposed halfbridge BDC. The design uses a half-bridge connected with the DC power supply on the primary side and a center-tapped transformer and a synchronous rectification on the secondary side. The converter can operate in two modes, namely, step-down mode and step-up mode. All of the four switches Q 1 –Q 4 are gated in both modes. Switch Q 1 is complementary with switch Q 4 , and switch Q 2 is complementary with switch Q 3 .
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Circuit diagram of the proposed converter
In the step-down mode, the primary side DC power supply VH (210–380 V) charges the secondary side battery VL (21–29 V), and Q 3 and Q 4 provide rectification. In contrast, Q 1 and Q 2 operate as rectifiers in the step-up mode when VL supplies the high side battery VH . For mathematical insight into the proposed converter, some assumptions are made as: (i) the ON-state resistance RDS(ON) of all switches is ignored; (ii) the capacitors C 1 , C 2 and C o are large enough, and the voltage across the capacitors can be taken as constant; and (iii) the capacitance of the capacitors C 1 and C 2 is equal i.e. C 1 =C 2 =C. Thus, V C1 = V C2 = VH /2; and (iv) D 1 –D 4 are the body diodes of Q 1 –Q 4 , and the diode forward resistance is zero.
- 2.2 Operational principles of the proposed BDC
- 2.2.1 Step-down mode
This is similar to a buck converter operational mode in which VH supplies VL with the charging current, iL . The equivalent circuits are shown in Fig. 3. The pulse-width modulation (PWM) technique is used to control the switches Q 1 –Q 4 . In both modes Q 1 and Q 2 are gated, with the duty cycle less than 0.5, while the duty cycle of Q 3 and Q 4 is more than 0.5. Fig. 4 shows some typical waveforms for the step-down mode. The operating principles during one switching period are described as follows:
Stage 1 [t0t1]: Switch Q 1 turns on and switch Q 4 turns off at t 0 , while switch Q 3 remains on. The current flow path for this stage is shown in Fig. 3(a) . In this stage v AB = VH /2, the current i 1 flows through Q 1 as i p , which is reflected from inductor current i L . The current i L increases linearly and flows totally through switch Q 3 to charge the battery VL
Stage 2 [t1t2]: Switch Q 1 turns off and switch Q 4 turns on at t 1 , while switch Q 3 still remains on. Because of the transformer leakage inductance l k1 , there is freewheeling current through D 2 [ Fig. 3(b) ], v CA = VH /2 and ip decreases linearly to zero. In this stage v A is clamped to ground so v Q2 = 0 and v Q1 = VH . Meanwhile i 4 increases and i 3 decreases linearly and at t 2 , i 3 = i 4 = iL /2.
Stage 3 [t2t3]: Switch Q 1 and Q 2 are in off state and v Q1 = v Q2 = VH /2. In this stage, v AB = 0 , no power is transferred to secondary side and the energy stored in the inductor L charges the low side battery VL [ Fig. 3(c) ]. The current iL is shared equally by switches Q 3 and Q 4 .
Stage 4 [t3t4]: Switch Q 2 turns on and switch Q 3 turns off at t 3 , while switch Q 4 remains on. The current flow path is shown in Fig. 3(d) . This is a similar operation to stage 1[ t 0 - t 1 ], but the voltage v AB =− VH /2. The current i 2 is built as − ip . In this stage, Q 4 is conducting and i 4 increases linearly as iL .
Stage 5 [t4t5]: Switch Q 2 turns off and switch Q 3 turns on at t 4 . Switch Q 4 still remains on. Because of l k1 , D 1 conducts and v A is clamped as VH , therefore v Q2 = VH and v Q1 =0 [ Fig. 3(e) ]. Meanwhile i 3 increases and i 4 decreases linearly and at t 3 , i 3 = i 4 = iL /2.
Stage 6 [t5t6]: The operation of this stage is the same as stage 3. The current path for this stage is shown in Fig. 3(f) .
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Equivalent circuits for the step-down mode
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Theoretical waveforms for the step-down mode
- 2.2.2 Step-up mode
For the step-up mode, the equivalent circuits considering the leakage inductance of the proposed converter are shown in Fig. 5 . In this operational mode, VL discharges to supply the primary side output voltage of VH with current i 1 or i 2 . Because of the existence of the transformer secondary side leakage inductance l k2 and l k3 , there will be current stress on Q 3 and Q 4 . For protection of Q 3 and Q 4 , the RC snubber circuit is necessary in parallel connection with Q 3 and Q 4 . The theoretical waveforms are shown in Fig. 6 , and modes of operation in one period ( t 0 t 6 ) are described as follows:
Stage 1 [t0t1]: Switch Q 3 is turned on at [ t 0 , with switch Q 4 remaining on while Q 1 and Q 2 are in the off state. The current flow path for this stage is shown in Fig. 5(a) . The secondary side of the transformer is effectively shorted, and v AB =0. Meanwhile, the energy is stored in the inductor L , while no energy is transferred to the primary side. In this stage, iL increases linearly and is divided equally between Q 3 and Q 4 . The primary side battery VH is charged by the capacitors C 1 and C 2 .
Stage 2 [t1t2]: Switch Q 4 is turned off while switch Q 1 is turned on at t 1 , with switch Q 3 remaining on. Because of the leakage inductance l k2 , there is stress on Q 4 and the RC snubber is charged by i 4 [ Fig. 5(b) ]. The current i 4 decreases linearly to zero and i 3 increases linearly to iL , building i1 as ip which increases linearly.
Stage 3 [t2t3]: In this stage v AB =0 [ Fig. 5(c) ], the energy stored in L is transferred to the primary side, iL and decrease in linearity. The capacitor C 2 is discharged and capacitor C 1 is charged.
Stage 4 [t3t4]: Switch Q 4 is turned on, with switch Q 3 remaining on while switch Q 1 is turned off at t 3 . This stage is similar to stage 1 in which the inductor L stores energy again, and the inductor current iL is equally shared by switches Q 3 and Q 4 [ Fig. 5(d) ]. Capacitor C 1 and C 2 discharge to supply the primary side DC source VH .
Stage 5 [t4t5]: At t4 Switch Q 3 is turned off, with switch Q 2 turned on and switch Q 4 remaining on. Because of the leakage inductance l k3 [ Fig. 5(e) ], there is stress on Q 3 and the RC snubber is charged by i 3 . The current i 3 decreases linearly to zero with i 4 increasing linearly to iL , building i 1 as − ip .
Stage 6 [t5t6]: In this stage [Fig. 5(f) ], i 4 is built as iL which decreases in linearity and energy is transferred to primary side. The capacitor C 2 is charged by i 2 conducted by Q 2 , which decreases linearly.
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Equivalent circuits for the step-up mode
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Theoretical waveforms of the step-up mode
3. Circuit Design Analysis
The BDC operates in step-down and step-up modes. Design parameters for the step-down mode will be discussed in detail. The parameters obtained can be used for step-up mode as well. The theoretical analysis and design guidelines will be discussed in this section.
- 3.1 Step-down mode
When the number of turns on the secondary windings are equal, that is, N 2 = N 3 , then N can be defined as the transformer turns ratio, as N 1 / N 2 or N 1 / N 3 . The relationship between VH and VL is expressed as:
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where D is the duty ratio of Q 1 and Q 2 .
To design the inductor L , the inductor current ripple Δ iL and the minimum duty ratio D min should be considered. The inductance L can be calculated as:
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When designing output capacitor C o , the transient overshoot should be taken into account. Because of the inductor L , the energy stored in L will be transferred to C o if there is a sudden change of load, causing a sudden change of V Co . According to the design specification, the overshoot voltage should be less than 3% of VL , so that for a 50% to full load situation, C o can be calculated as:
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and for the equivalent series resistance (ESR) of C o , the value of ESR should be limited by the following equation:
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So several capacitors may be connected in parallel, if necessary, to meet the requirements of ESR.
The two capacitors C 1 and C 2 should be large enough to constrain the input current ripple and equally share VH for Q 1 and Q 2 .
In practical applications, there are many reasons causing the voltage imbalance. A few of those are: 1) the conduction periods of the two high side switches Q 1 and Q 2 are not strictly equal, 2) C 1 and C 2 are charged and discharged in turns, and if the capacitance value is not big enough, there will be big voltage ripple which may cause voltage imbalance.
To avoid the voltage imbalance, it is necessary to keep the equal conduction periods for Q 1 and Q 2 to the maximum possibility, and it should be noted that the switch Q 4 is complementary with Q 1 and it is the same situation for Q 2 and Q 3 . For C 1 and C 2 , there is
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where ip is the transformer primary side current.
It can be seen that if C is big enough, the ripple will be small and it will not affect the circuit operations.
The maximum voltage stress and RMS/max current ratings should be considered when selecting the switches of both sides. Q 1 and Q 2 have the ratings of:
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Q 3 and Q 4 have the ratings of:
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- 3.2 Step-up mode
For this mode, VL is the input voltage, and VH is output voltage. Therefore:
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where D ’ is the duty ratio of Q 3 and Q 4 .
Because the BDC operates just as the current flows inversely, but the voltage polarity remains unchanged, so the design parameters of all components of step-down mode can be employed in the step-up mode.
4. Experimental Results
A 3 kW prototype, as shown in Fig. 7 , was built and tested to evaluate the performance of the proposed BDC. The experimental parameters and circuit components are given in Table 1 . The experimental waveforms of stepdown and step-up modes are shown in Fig. 8 to Fig. 13 .
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Detailed photograph of the 3 kW proposed BDC
Experimental parameters
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Experimental parameters
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Step-down mode experimental waveforms at light load
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Step-down mode experimental waveforms at full load
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Step-down mode experimental waveforms at 1 kW
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Step-up mode experimental waveforms at light load
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Step-up mode experimental waveforms at full load
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Step-up mode experimental waveforms at 2 kW
Fig. 8 shows the voltage across two switches Q 1 and Q 4 , which is V Q1 and V Q4 , respectively, and the input and the output voltage VH of 380 V and VL of 29.6 V at light load. One can see that the voltage stress on switch Q 1 is equal to VH , and that on switch Q 4 is equal to VH / N . The output voltage VL is well regulated at 29 V. In this situation, the duty ratio of switches Q 1 and Q 2 is 23.4%.
Fig. 9 shows the inductor current iL and the switch voltages V Q1 and V Q4 at full load. It can be seen that the average output current iL,Avg is 107.75 A, achieving 3.1 kW output power.
Fig. 10 shows the step-down 1 kW experimental waveforms. Channel 1 and channel 3 show the drain-to-source voltage V Q3 and V Q2 . The input voltage can be seen from the maximum value of VQ2(max) as 377 V and the output voltage is 29 V. In this situation the duty ratio of Q 1 and Q 2 is about 23.1%. Channel 2 shows transformer primary side current which changes the direction accordingly when Q 1 or Q 2 are turned on. The average value of every conduction period is about 13 A. Channel 4 shows the voltage across L .
Fig. 11 shows V Q1 , V Q4 , VL and VH for the step-up mode at light loads. It should be noted that the output voltage is well regulated at 380 V, and the maximum voltage on switch Q 1 is equal to VH and that on switch Q 4 is VH /N. In this situation, the duty ratio of Q 3 and Q 4 is 77.2%.
Fig. 12 shows the average input current of −108.08 A, output current of 7.26 A, and the voltage across switches Q 1 and Q 4 at full load conditions. The input and output power obtained are 3.13 kW and 2.76 kW, respectively.
Fig. 13 shows the step up mode 2 kW experimental waveforms. The input voltage is 29 V and the output voltage can be seen from VQ2(max) as 351 V. In this situation the duty ratio of Q3 and Q4 is about 75.2%. Channel 2 shows transformer primary side current iP with the average value of every conduction period about 23 A. Channel 4 shows the voltage across the inductor L .
Fig. 14 shows the measured efficiencies of the proposed BDC when it operates in step-down mode. The converter achieves efficiency of higher than 90% from 30% to full load.
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System efficiency in step-down mode
Fig. 15 shows the measured efficiencies of the proposed converter when it operates in step-up mode. It can be seen that more than 88% of the efficiency is achieved from 30% to full load.
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System efficiency in step-up mode
5. Conclusion
A 3 kW BDC for EVs is proposed in this paper. With a simple topology structure, the converter consists of a half-bridge on the primary side and a synchronous rectifier on the secondary side. Four switches are employed in both the step-down mode and step-up mode. The theoretical analyses have been proved by the experimental results of the 3 kW prototype circuit. When operating in step-down mode, an efficiency was achieved of more than 90% from 30% to the full load, while an efficiency of 88.2% is achieved at full load for the step-up mode.
Acknowledgements
This work was supported by the Human Resources Program in Energy Technology of the Korea Institute of Energy Technology Evaluation and Planning (KETEP), granted financial resources from the Ministry of Trade, Industry and Energy, Republic of Korea. (20154030200730)This work was supported by the Energy Efficiency & Resources Department of the Korea Institute of Energy Technology Evaluation and Planning (KETEP) and was granted financial resources from the Ministry of Trade, Industry and Energy, Republic of Korea. (20132010101950)
BIO
Arsalan Ansari received his B.E. degree in electrical engineering from the Mehran University of Engineering and Technology, Jamshoro, Pakistan, in 2011. He is currently moving towards the completion of his Ph.D. degree in electrical engineering at Hanyang University, Ansan, Korea. He was appointed as a Lecturer with Mehran University of Engineering and Technology. His current research interests include bidirectional DC-DC converters, multilevel inverters and grid-connected renewable energy systems. Mr. Ansari was a recipient of the Scholarship for the Integrated Master and Ph.D. Program by the Government of Pakistan. He was a medalist among his engineering batch.
Puyang Cheng received his B.S. degree in Mechanical Engineering and Automation from Wuhan University of Science and Technology, Wuhan, China, in 2013. He is currently working toward the M.S. degree in Electrical Engineering in Hanyang University, Ansan, Korea. His research interests are bidirectional DC/DC converters and power factor correction.
Hee-Jun Kim received the B.S and M.S. degree in electronics engineering from Hanyang University, Seoul, Korea, in 1976 and 1978, respectively, and the Ph.D. degree from Kyushu University, Fukuoka, Japan, in 1986, all in electronics engineering. Since 1987, he has been a Professor with Hanyang University, Ansan, Korea. His current research interests include switching power converters, soft-switching techniques, and analog signal processing. Prof. Kim is a senior member of IEEE.
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